Control of a brushless motor

ABSTRACT

A method of controlling a brushless motor that includes exciting a winding of the motor in advance of predetermined rotor positions by an advance period. The length of the advance period is defined by a waveform that varies periodically with time. Additionally, a control system that implements the method, and a motor system that incorporates the control system.

REFERENCE TO RELATED APPLICATIONS

This application claims the priority of United Kingdom Application No.1006386.5, filed Apr. 16, 2010, the entire contents of which areincorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates to control of a brushless motor.

BACKGROUND OF THE INVENTION

A brushless motor typically includes a control system for controllingthe excitation of phase windings of the motor. When driven by an ACsupply, the control system often includes an active power factorcorrection (PFC) circuit, which outputs a regular DC voltage for use inexciting the phase windings while ensuring that current drawn from theAC supply is substantially sinusoidal. As a result, a relatively highpower factor may be achieved. However, an active PFC circuitsignificantly increases the cost of the control system. Additionally,the PFC circuit requires a high-capacitance DC link capacitor, which isboth physically large and expensive, in order to provide a regularfeedback voltage to the PFC circuit.

SUMMARY OF THE INVENTION

In a first aspect, the present invention provides a method ofcontrolling a brushless motor, the method comprising exciting a windingof the motor in advance of predetermined rotor positions by an advanceperiod, the advance period being defined by a waveform that variesperiodically with time.

The length of the advance period influences the amount of current driveninto the winding over an electrical half-cycle of the motor, which inturn influences the amount of current drawn from the power supply usedto excite the winding. The waveform of the advance period may thereforebe defined so as to shape the waveform of current drawn from the powersupply. In particular, the waveform of the advance period may be definedso as to reduce the magnitude of low-order harmonics within the currentwaveform. Consequently, an increase in power factor may be achieved.

The advance period and the conduction period (i.e. the period over whichthe winding is excited during each electrical half-cycle) may togetherbe defined such that the waveform of current drawn from the power supplyapproaches that of a sinusoid. Consequently, a relatively high powerfactor may be achieved without the need for a PFC circuit orhigh-capacitance link capacitor.

The length of the conduction period influences the magnitude of themagnetic flux density in the motor. The conduction period may thereforevary with time so as to achieve a particular envelope for the magneticflux density. In particular, the waveform of the conduction period maybe defined so as to reduce the peak magnetic flux density in the motor.The waveform of the advance period may then be defined so as tocompensate for any loss in power that arises from the variation in theconduction period. Consequently, a more efficient and/or smaller motormay be achieved.

The method may comprise determining an advance period for eachelectrical half-cycle of the motor. This then helps achieve a smootherenvelope for the current waveform drawn from the power supply. It is notnecessary that a different advance period is used for successiveelectrical half-cycles of the motor. Depending on the shape of thewaveform for the advance period as well as the length of each electricalhalf-cycle, it is quite possible for successive electrical half-cyclesof the motor to have the same advance period. For example, if thewaveform of the advance period were to include a flat spot, then it isquite possible for two successive electrical half-cycles of the motor tohave the same advance period.

For a permanent-magnet motor, the variation in the waveform of theadvance period may take the form of an inverted triangle, invertedtrapezoid or inverted half-sinusoid over each cycle of the waveform. Theadvance period therefore decreases over the first half of each cycle ofthe waveform and increases over the second half of each cycle. Each ofthese waveforms has been found to work well in reducing the magnitude oflow-order harmonics within the current waveform drawn from the powersupply.

For a reluctance motor, the conduction period may be defined by awaveform that decreases over the first half of each cycle of thewaveform and increases over the second half of each cycle. This waveformhas been found to work well in controlling the level of magnetic flux inthe motor when the winding is excited with a voltage having a relativelyhigh ripple. In particular, by employing a conduction period that has aminimum in the region of maximum voltage, the peak magnetic flux densityin the motor may be reduced. The advance period may then be defined by awaveform that increases over the first half of each cycle of thewaveform and decreases over the second half of the waveform. Forexample, the variation in the waveform of the advance period may takethe form of a triangle, trapezoid or half-sinusoid over each cycle ofthe waveform. Consequently, any loss in power that may arise from thevariation in the conduction period may be compensated by variation inthe advance period.

The method may comprise adjusting the waveform of the advance period inresponse to a change in one of motor speed and excitation voltage. As aresult, a particular power profile may be achieved over a range of motorspeeds and/or voltages. In particular, by appropriate adjustment of thewaveform, constant average power may be achieved over a range of motorspeeds and/or voltages. Changes in the motor speed and/or the excitationvoltage may also increase the magnitude of low-order harmonics withinthe current waveform drawn from the power supply. By adjusting thewaveform in response to changes in speed and/or voltage, the magnitudeof low-order harmonics may be kept relatively small over a range ofspeeds and/or voltages.

The length of the advance period may comprise the sum of a firstcomponent and a second component. The first component is then constantover each cycle of the waveform, while the second component varies overeach cycle of the waveform. The first component thus acts as an offsetor uplift for the waveform of the advance period. Consequently, a higheraverage input power may be achieved for a given peak current. The secondcomponent may vary as a half-sinusoid over each cycle of the waveform.Consequently, a current waveform having relatively small low-orderharmonics may be achieved.

Defining the advance period as the sum of two components provides aconvenient method of adjusting the waveform in response to changes inmotor speed and/or excitation voltage. In particular, the method mayinvolve adjusting the first component only in response to a change inmotor speed and/or excitation voltage.

The method may involve rectifying an alternating voltage to provide arectified voltage, and exciting the winding with the rectified voltage.The length of the advance period may then vary over each half-cycle ofthe alternating voltage. More particularly, the waveform of the advanceperiod may repeat with each half-cycle of the alternating voltage. As aresult, a current waveform having reduced low-order harmonics may beachieved that repeats with each half-cycle of the alternating voltage.Improvements in power factor may therefore be achieved without the needfor a PFC circuit or high-capacitance link capacitor.

By exciting the winding in advance of unaligned rotor positions, currentis initially driven into the winding during a period of negative torque.However, by driving current into the winding at an earlier stage, morecurrent may be driven into the winding during the period of positivetorque. Consequently, the negative torque is more than offset by thesubsequent gain in positive torque. As the alternating voltage increasesand decreases, the amount of current that may be driven into the windingover each electrical half-cycle of the motor also increases anddecreases. As the alternating voltage increases, the same amount oftorque may therefore be generated using a smaller advance period.Importantly, by employing a smaller advance period, the amount ofnegative torque is reduced. Consequently, by varying the advance periodacross each half-cycle of the alternating voltage, a more efficientmotor system may be realized. To this end, the advance period maydecrease over a first half of each half-cycle of the alternating voltageand increase over a second half of each half-cycle of the alternatingvoltage.

The length of the advance period may be defined by the length of timethat has elapsed since a zero-crossing in the alternating voltage. Forexample, the method may involve measuring the length of time that haselapsed since a zero-crossing in the alternating voltage and thendetermining the advance period using the measured time. This thenensures that the waveform of the advance period is synchronized with thewaveform of the alternating voltage.

The length of the advance period may comprise the sum of a firstcomponent and a second component. The first component is then constantover each half-cycle of the alternating voltage, and the secondcomponent varies over each half-cycle of the alternating voltage. Asnoted above, the first component acts as an offset or uplift for thewaveform of the advance period. Consequently, a higher average inputpower may be achieved for a given peak current. The second component maybe determined using the length of time that has elapsed since azero-crossing in the alternating voltage. Consequently, as noted above,the waveform of the advance period is then synchronized with thewaveform of the alternating voltage.

The method may comprise adjusting the first component in response tochanges in the motor speed and/or the RMS value of the alternatingvoltage. A particular power profile may therefore be achieved over arange of motor speeds and/or voltages. Additionally or alternatively, acurrent waveform having relatively small low-order harmonics may beachieved over a range of motor speeds and/or voltages. The method mayalso comprise adjusting the time used to determine the second componentin response to changes in motor speed and/or RMS voltage. By adjustingthe time used to determine the second component, the phase of thewaveform of the advance period relative to the alternating voltage isadjusted. Adjusting the phase of the waveform of the advance period mayprove useful in maintaining relatively small low-order harmonics over arange of motor speeds and/or voltages.

The waveform of the advance period may be stored as one or more lookuptables. For example, the method may comprise storing a first lookuptable of first control values. The first component is then obtained byindexing the first lookup table using speed and/or voltage to select afirst control value, and using the first control value to determine thefirst component. The method may also comprise storing a second lookuptable of second control values. The second component is then obtained byindexing the second lookup table using the time that has elapsed since azero-crossing in the alternating voltage to select a second controlvalue, and using the selected second control value to determine thesecond component. The use of lookup tables simplifies the control of themotor. Consequently, a relatively simple and cheap controller may beused to implement the method. Each control value may be an absolutevalue or a difference value. Where the control values are differencevalues, the method also comprises storing a reference value to which thedifference value is applied to obtain the first or second component.Storing a difference value often requires less memory than an absolutevalue. As a result, the lookup tables may be stored more efficiently.

The method may also comprise adjusting the time used to index the secondlookup table in response to changes in motor speed and/or RMS voltage.By adjusting the time used to index the second lookup table, the phaseof the waveform of the advance period relative to the waveform of thealternating voltage is adjusted. The method may comprise storing a thirdlookup table of third control values. The third lookup table is thenindexed using speed and/or voltage to select a third control value,which is then used to adjust the time used to index the second lookuptable. The use of a lookup table to adjust the phase of the waveform ofthe advance period simplifies the control of the motor.

In a second aspect, the present invention provides a control system fora brushless motor, the control system performing the method described inany one of the preceding paragraphs.

The control system may comprise a position sensor for sensing theposition of a rotor of the motor, and a controller for generating one ormore control signals for exciting the winding in advance of thepredetermined rotor positions by the advance period. The controller thendetermines the advance period and generates the control signals inresponse to each edge of a signal output by the position sensor.Consequently, the controller determines an advance period for eachelectrical half-cycle of the motor. This then helps achieve a smootherenvelope for the current waveform drawn from the power supply.

The control system may comprise a rectifier that rectifies analternating voltage to provide a rectified voltage, a zero-crossdetector for detecting zero-crossings in the alternating voltage, aninverter coupled to the winding, and a controller for controlling theinverter. The controller uses a signal output by the zero-cross detectorto measure the time that has elapsed since a zero-crossing in thealternating voltage. From this measurement, the controller determinesthe advance period to be used for each electrical half-cycle of themotor. The controller then generates one or more control signals forexciting the winding in advance of the predetermined rotor positions bythe advance period, and the inverter excites the winding with therectified voltage in response to the control signals.

In a third aspect, the present invention provides a motor systemcomprising a permanent-magnet motor and a control system as described inany one of the preceding paragraphs.

The motor therefore comprises a permanent-magnet rotor, which induces aback EMF in the winding of the motor. The back EMF makes it difficult toaccurately control the amount of current drawn from the power supply.With the control employed by the motor system, the winding is excited inadvance of zero-crossing of back EMF by an advance period that variesperiodically with time. Through the use of an appropriate waveform forthe advance period, a current waveform having improved low-orderharmonics may be achieved.

BRIEF DESCRIPTION OF THE DRAWINGS

In order that the present invention may be more readily understood,embodiments of the invention will now be described, by way of example,with reference to the accompanying drawings, in which:

FIG. 1 is a block diagram of a motor system in accordance with thepresent invention;

FIG. 2 is a schematic diagram of the motor system;

FIG. 3 is a sectional view of the motor of the motor system;

FIG. 4 details the allowed states of the inverter in response to controlsignals issued by the controller of the motor system;

FIG. 5 is a schematic diagram of the current regulator of the motorsystem;

FIG. 6 illustrates the overrun period used by the controller whenoperating in single-switch mode;

FIG. 7 illustrates a three-step process used by the controller whenmeasuring analog input signals;

FIG. 8 details the various operating modes of the motor system;

FIG. 9 details the direction in which the motor is driven in response tocontrol signals issued by the controller;

FIG. 10 illustrates various waveforms of the motor system when operatingin Low-Speed Acceleration Mode;

FIG. 11 illustrates various waveforms of the motor system when operatingin High-Speed Acceleration Mode;

FIG. 12 illustrates various waveforms of the motor system when operatingin Running Mode;

FIG. 13 illustrates the current waveform drawn from the power supply ofthe motor system when operating in Running Mode;

FIG. 14 illustrates various waveforms and interrupts of the motor systemwhen operating in overcurrent single-switch mode;

FIG. 15 illustrates various waveforms and interrupts of the motor systemwhen operating in unlimited-freewheel single-switch mode;

FIG. 16 is a schematic diagram of a timer and comparator module arrangedto generate a control signal;

FIG. 17 is a schematic diagram of a timer and PWM module arranged togenerate a control signal;

FIG. 18 illustrates various waveforms and interrupts of the motor systemwhen operating in limited-freewheel single-switch mode;

FIG. 19 details the values of various hardware components for aparticular embodiment of motor system in accordance with the presentinvention;

FIG. 20 details various constants and thresholds employed by thecontroller of the particular motor system;

FIG. 21 illustrates the flux-linkage characteristics of a link inductorof the particular motor system;

FIG. 22 illustrates the flux-linkage characteristics of the motor of theparticular motor system;

FIG. 23 details the various operating modes of the particular motorsystem;

FIG. 24 details a map of control values used by the controller of theparticular motor system when operating in multi-switch mode;

FIG. 25 details a map of control values used by the controller of theparticular motor system when operating in overcurrent single-switchmode;

FIG. 26 details a portion of the advance lookup table used by thecontroller of the particular motor system when operating inunlimited-freewheel single-switch mode;

FIG. 27 details a portion of the offset lookup table used by thecontroller of the particular motor system when operating inunlimited-freewheel single-switch mode;

FIG. 28 details a portion of the phase lookup table used by thecontroller of the particular motor system when operating inunlimited-freewheel single-switch mode;

FIG. 29 details a portion of the sine map used by the controller of theparticular motor system when operating in single-switch mode;

FIG. 30 illustrates possible waveforms for the conduction period used bythe controller in single-switch mode; and

FIG. 31 illustrates possible waveforms for the advance period of analternative motor system in accordance with the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The motor system 1 of FIGS. 1 to 3 comprises a brushless motor 2 and acontrol system 3. Power to the motor system 1 is provided by an ACsupply 4. The AC supply 4 is intended to be a domestic mains supply,though other power supplies capable of providing an alternating voltagemight equally be used.

The motor 2 comprises a four-pole permanent-magnet rotor 5 that rotatesrelative to a stator 6. The stator 6 comprises a pair of c-shaped coresthat define four stator poles. Conductive wires are wound about thestator 6 and are coupled together to form a single phase winding 7.

The control system 3 comprises a rectifier 8, a DC link filter 9, aninverter 10, a gate driver module 11, a current sensor 12, a positionsensor 13, a zero-cross detector 14, a temperature sensor 15, and acontroller 16.

The rectifier 8 is a full-wave bridge D1-D4 that rectifies the output ofthe AC supply 4 to provide a DC voltage.

The DC link filter 9 comprises a link capacitor C1 and a link inductorL1. The link capacitor C1 acts to smooth the relatively high-frequencyripple that arises from inverter switching. As described below in moredetail, the link capacitor C1 is not required to smooth the rectified DCvoltage at the fundamental frequency. Consequently, a link capacitor ofrelatively low capacitance may be used. The link inductor L1 acts tosmooth any residual current ripple that arises from inverter switching.Again, since the link inductor L1 is intended to reduce ripple at theswitching frequency of the inverter 10, an inductor of relatively lowinductance may be used. In order that saturation is avoided, the linkinductor L1 has a saturation point that exceeds the peak current drawnfrom the AC supply 4 during normal operation of the motor system 1.

The inverter 10 comprises a full bridge of four power switches Q1-Q4that couple the DC link voltage to the phase winding 7. Each powerswitch Q1-Q4 is an IGBT, which is capable of operating at the voltagelevel typically of most mains power supplies. Other types of powerswitch, such as BJTs or MOSFETs, might alternatively be used dependingon the rating of the power switch and the voltage of the AC supply 4.Each of the switches Q1-Q4 includes a flyback diode, which protects theswitch against voltage spikes that arise during inverter switching.

The gate driver module 11 drives the opening and closing of the switchesQ1-Q4 of the inverter 10 in response to control signals received fromthe controller 16.

The current sensor 12 comprises a pair of shunt resistors R1,R2, eachresistor located on a lower arm of the inverter 10. The resistance ofeach shunt resistor R1,R2 is ideally as high as possible withoutexceeding dissipation limits during normal operation of the motor system1. The voltage across each shunt resistor R1,R2 is output to thecontroller 16 as a current sense signal, I_SENSE_1 and I_SENSE_2. Thefirst current sense signal, I_SENSE_1, provides a measure of the currentin the phase winding 7 when driven from right to left (as is describedbelow in more detail). The second current sense signal, I_SENSE_2,provides a measure of the current in the phase winding 7 when drivenfrom left to right. In locating the shunt resistors R1,R2 on the lowerarms of the inverter 10, current in the phase winding 7 continues to besensed during freewheeling (again, as is described below in moredetail).

The position sensor 13 is a Hall-effect sensor that outputs a digitalsignal, HALL, that is logically high or low depending on the directionof magnetic flux through the sensor 13. By locating the position sensor13 adjacent the rotor 5, the HALL signal provides a measure of theangular position of the rotor 5. More particularly, each edge of theHALL signal indicates a change in the polarity of the rotor 5. Whenrotating, the permanent-magnet rotor induces a back EMF in the phasewinding 7. Consequently, each edge of the HALL signal also represents achange in the polarity of the back EMF in the phase winding 7.

The zero-cross detector 14 comprises a pair of clamping diodes D5,D6that output a digital signal, Z_CROSS, that is logically high when thevoltage of the AC supply 4 is positive and logically low when thevoltage of the AC supply 4 is negative. Each edge of the Z_CROSS signalthus represents a point in time at which the voltage of the AC supply 4crosses zero.

The temperature sensor 15 comprises a thermistor R7 that outputs ananalog signal, TEMP, that provides a measure of the temperature withinthe motor system 1.

The controller 16 comprises a microcontroller having a processor 17, amemory device 18, a plurality of peripherals 19 (e.g. ADC, comparators,timers etc.), a plurality of input pins 20, and a plurality of outputpins 21. The memory device 18 stores software instructions for executionby the processor 17. The memory device 18 also stores a plurality oflookup tables, which are indexed by the processor 17 during operation ofthe motor system 1.

The controller 16 is responsible for controlling the operation of themotor system 1. In response to signals at the input pins 20, thecontroller 16 generates control signals at the output pins 21. Theoutput pins 21 are coupled to the gate drive module 11, which controlsthe opening and closing of the switches Q1-Q4 of the inverter 10 inresponse to the control signals.

Seven signals are received at the input pins 20 of the controller 16:I_SENSE_1, I_SENSE_2, HALL, Z_CROSS, TEMP, DC_LINK and DC_SMOOTH.I_SENSE_1 and I_SENSE_2 are the signals output by the current sensor 12.HALL is the signal output by the position sensor 13. Z_CROSS is thesignal output by the zero-cross detector 14. TEMP is the signal outputby the temperature sensor 15. DC_LINK is a scaled-down measure of the DClink voltage, which is obtained by a potential divider R3,R4 locatedbetween the DC link line and the zero volt line. DC_SMOOTH is a smoothedmeasure of the DC link voltage, obtained by a potential divider R5,R6and smoothing capacitor C2.

In response to the signals received at the inputs, the controller 16generates and outputs four control signals: TRIP#, DIR1, DIR2, andFREEWHEEL#.

TRIP# is a failsafe control signal. When TRIP# is pulled logically low,the gate driver module 11 opens all switches Q1-Q4 of the inverter 10.As described below in more detail, the controller 16 pulls TRIP#logically low in the event that current through the phase winding 7exceeds a failsafe threshold.

DIR1 and DIR2 control the direction of current through the inverter 10and thus through the phase winding 7. When DIR1 is pulled logically highand DIR2 is pulled logically low, the gate driver module 11 closesswitches Q1 and Q4, and opens switches Q2 and Q3, thus causing currentto be driven through the phase winding 7 from left to right. Conversely,when DIR2 is pulled logically high and DIR1 is pulled logically low, thegate driver module 11 closes switches Q2 and Q3, and opens switches Q1and Q4, thus causing current to be driven through the phase winding 7from right to left. Current in the phase winding 7 is thereforecommutated by reversing DIR1 and DIR2. If both DIR1 and DIR2 are pulledlogically low, the gate drive module 11 opens all switches Q1-Q4.

FREEWHEEL# is used to disconnect the phase winding 7 from the DC linkvoltage and allow current in the phase winding 7 to re-circulate orfreewheel around the low-side loop of the inverter 10. Accordingly, inresponse to a FREEWHEEL# signal that is pulled logically low, the gatedriver module 11 causes both high-side Q1,Q2 switches to open. Currentthen freewheels around the low-side loop of the inverter 10 in adirection defined by DIR1 and DIR2.

FIG. 4 summarizes the allowed states of the switches Q1-Q4 in responseto the control signals of the controller 16. Hereafter, the terms ‘set’and ‘clear’ will be used to indicate that a signal has been pulledlogically high and low respectively.

When a particular control signal changes, there is a short delay betweenthe changing of the control signal and the physical opening or closingof a power switch. If a further control signal were changed during thisdelay period, it is possible that both switches on a particular arm ofthe inverter (i.e. Q1,Q3 or Q2,Q4) may be closed at the same time. Thisshort-circuit, or shoot-through as it is often termed, would damage theswitches on that particular arm of the inverter 10. Accordingly, inorder to prevent shoot-through, the controller 16 employs a dead time,T_DT, between the changing of two control signals. So, for example, whencommutating the phase winding 7, the controller 16 first clears DIR1,waits for the dead time, T_DT, and then sets DIR2. The dead time isideally kept as short as possible so as to optimize performance whileensuring that the gate driver module 11 and the power switches Q1-Q4have sufficient time to respond.

Commutation

The controller 16 commutates the phase winding 7 in response to edges ofthe HALL signal. Commutation involves reversing DIR1 and DIR2 (i.e.clearing DIR1 and setting DIR2, or clearing DIR2 and setting DIR1) so asto reverse the direction of current through the phase winding 7. Thephase winding 7 may be freewheeling at the point of commutation.Accordingly, in addition to reversing DIR1 and DIR2, the controller 16sets FREEHWEEL#.

Synchronous Commutation

Below a predetermined speed threshold, SPEED_ADV, the controller 16commutates the phase winding 7 in synchrony with the edges of the HALLsignal. Each edge of the HALL signal represents a change in the polarityof back EMF in the phase winding 7. Consequently, at speeds belowSPEED_ADV, the controller 16 commutates the phase winding 7 in synchronywith the zero-crossings of back EMF.

As the rotor 5 accelerates, the period of each electrical half-cycledecreases and thus the time constant (L/R) associated with theinductance of the phase winding 7 becomes increasingly important.Additionally, the magnitude of back EMF in the phase winding 7increases, which then influences the rate at which current rises in thephase winding 7. Consequently, if the controller 16 continued tocommutate the phase winding 7 in synchrony with the edges of the HALLsignal, a speed would be reached at which it would be no longer possibleto drive additional current into the phase winding 7 over eachelectrical half-cycle. Accordingly, on reaching SPEED_ADV, thecontroller 16 switches from synchronous commutation to advancedcommutation. By commutating the phase winding 7 in advance of the edgesof the HALL signal, the voltage used to excite the phase winding 7 isboosted by the back EMF. As a result, the direction of current throughthe phase winding 7 may be more quickly reversed. Additionally, phasecurrent may be caused to lead the back EMF, which then helps tocompensate for the slower rate of current rise. Although this thengenerates a short period of negative torque, this is normally more thancompensated by the subsequent gain in positive torque.

Advanced Commutation

At speeds at or above the speed threshold, SPEED_ADV, the controller 16commutates the phase winding 7 in advance of each edge of the HALLsignal by an advance period, T_ADV. Since the electrical half-cycleperiod decreases and the back EMF increases with rotor speed, theelectrical angle at which commutation occurs in advance of the edges ofthe HALL signal ideally increases with rotor speed. For a particularadvance period, T_ADV, the corresponding advance angle, A_ADV, may bedefined as:

A_ADV(elec. deg)=T_ADV(sec)*{ω(rpm)/60}*360(mech. deg)*n/2

where A_ADV is the advance angle in electrical degrees, T_ADV is theadvance period in seconds, ω is the rotor speed in rpm, and n is thenumber of rotor poles. From this equation, it can be seen that theadvance angle is directly proportional to the rotor speed. Consequently,even for a fixed advance period, the advance angle increases with rotorspeed. However, better control over acceleration, power and efficiencymay be achieved by employing different advance periods at differentrotor speeds. The controller 16 therefore comprises an advance lookuptable that stores an advance period for each of a plurality of rotorspeeds.

In response to an edge of the Z_CROSS signal, the controller 16 selectsfrom the advance lookup table an advance period, T_ADV, corresponding tothe speed of the rotor 5. The speed of the rotor 5 is determined fromthe interval, T_HALL, between two successive edges of the HALL signal.This interval will hereafter be referred to as the Hall period. Thespeed of the rotor 5 is then defined by:

ω(rpm)=60/{n*T_HALL(sec)}

where ω is the speed of the rotor in rpm, T_HALL is the Hall period inseconds, and n is the number of poles of the rotor. The controller 16uses the selected advance period to commutate the phase winding 7 inadvance of the edges of the HALL signal. The same advance period, T_ADV,is then used by the controller 16 until such time as a further edge ofthe Z_CROSS signal is detected. In response to the further edge of theZ_CROSS signal, the controller 16 selects from the advance lookup tablea new advance period corresponding to the speed of the rotor 5. Theadvance period is therefore updated only when the voltage of the ACsupply 4 crosses zero, and is constant over each half-cycle of the ACsupply 4.

In order to commutate the phase winding 7 in advance of a particularedge of the HALL signal, the controller 16 acts in response to thepreceding edge of the HALL signal. In response to an edge of the HALLsignal, the controller 16 subtracts the advance period, T_ADV, from theHall period, T_HALL, in order to obtain a commutation period, T_COM:

T_COM=T_HALL−T_ADV

The controller 16 then commutates the phase winding 7 at a time, T_COM,after the edge of the HALL signal. As a result, the phase winding 7 iscommutated in advance of the subsequent edge of the HALL signal by theadvance period, T_ADV.

As noted above, the advance period, T_ADV, remains fixed over eachhalf-cycle of the AC supply 4. However, the speed of the rotor 5 variesover each half-cycle of the AC supply 4 owing to the sinusoidal increaseand decrease in DC link voltage. The Hall period, T_HALL, thereforevaries over each half-cycle of the AC supply 4. Consequently, incontrast to the advance period, the controller 16 calculates thecommutation period, T_COM, for each edge of the HALL signal.

Current Control

A number of the peripherals 19 of the controller 16 are configured todefine a current regulator 22. The current regulator 22 monitors andregulates current in the phase winding 7. The current regulator 22performs two functions. First, the current regulator 22 clears TRIP# inthe event that current in the phase winding 7 exceeds a failsafethreshold. Second, the current regulator 22 generates an overcurrentsignal in the event that current in the phase winding 7 exceeds anovercurrent threshold.

As illustrated in FIG. 5, the current regulator 22 comprises a failsafemodule 23 and an overcurrent module 24.

The failsafe module 23 comprises a multiplexer 25, a comparator 26, aNOT gate 27, and an SR latch 28. The multiplexer 25 has two inputs forselecting one of the two current-sense signals, I_SENSE _1 andI_SENSE_2. The selection made by the multiplexer 25 is controlled by theprocessor 17 in response to the direction of current through the phasewinding 7. In particular, when DIR1 is set, the multiplexer 25 is causedto select I_SENSE_1, and when DIR2 is set, the multiplexer is caused toselect I_SENSE_2. The output of the multiplexer 25 is delivered to thecomparator 26, which compares the voltage of the selected current-sensesignal against a predetermined failsafe voltage, TRIP_REF. TRIP_REF isset such that the output of the comparator 26 is pulled logically highwhen current through the selected shunt resistor R1,R2 is greater than apredetermined failsafe threshold, I_MAX. TRIP_REF is thus defined byI_MAX and the resistances of the shunt resistors R1,R2. The output ofthe comparator 26 is delivered to the NOT gate 27, the output of whichis delivered to the S-input of the SR latch 28. The Q# output of the SRlatch 28 is output by the current regulator 22 as the TRIP# signal.Consequently, when the voltage of the current-sense signal, I_SENSE_1 orI_SENSE_2, is greater than TRIP_REF, TRIP# is cleared.

As noted above, the gate driver module 11 opens all switches Q1-Q4 ofthe inverter 10 in response to a cleared TRIP# signal. The failsafemodule 23 of the current regulator 22 thus prevents current in the phasewinding 7 from exceeding a failsafe threshold, I_MAX, above which theswitches Q1-Q4 may be damaged and/or the rotor 5 may be demagnetized. Byemploying hardware to clear the TRIP# signal, the current regulator 22responds relatively quickly when current in the phase winding 7 exceedsthe failsafe threshold. If software executed by the processor 17 wereinstead employed to clear the TRIP# signal, a delay might arise betweencurrent exceeding the failsafe threshold and the clearing of the TRIP#signal, during which time current may rise to a level that damages theswitches Q1-Q4 or demagnetizes the rotor 5.

The processor 17 polls the TRIP# signal in response to each edge of theHALL signal. If the TRIP# signal is clear for five successive HALLedges, the processor 17 writes a ‘Failsafe Exceeded’ error to the memorydevice 18 and enters Fault Mode, which is described below in moredetail. Monitoring the TRIP# signal in this manner ensures that thecontroller 16 does not inadvertently enter Fault Mode due to transientnoise in the TRIP# signal.

The overcurrent module 24 comprises a multiplexer 29 and a comparator30. The multiplexer 29, like that of the failsafe module 23, has twoinputs for selecting one of the two current-sense signals, I_SENSE_1 andI_SENSE_2. Again, the selection made by the multiplexer 29 is controlledby the processor 17 in response to the direction of current through thephase winding 7. Consequently, when DIR1 is set, the multiplexer 29selects I_SENSE_1, and when DIR2 is set, the multiplexer 29 selectsI_SENSE_2. The output of the multiplexer 29 is delivered to thecomparator 30, which compares the voltage of the current-sense signalagainst the voltage of the DC_LINK signal. When the current-sensesignal, I_SENSE_1 or I_SENSE_2, is greater than DC_LINK, the output ofthe comparator 30 is pulled logically low. The overcurrent module 24thus outputs an overcurrent signal that is pulled logically low whencurrent in the phase winding 7 exceeds an overcurrent threshold that isproportional to the DC link voltage.

The output of the overcurrent module 24 is coupled to the processor 17,which executes an overcurrent routine in response to a low overcurrentsignal. Since the overcurrent threshold is proportional to the DC linkvoltage, the overcurrent threshold varies as a rectified sinusoid acrosseach cycle of the AC supply 4, the benefits of which are explained inmore detail below.

The resistances of the potential divider R3,R4 are selected such thatthe peak voltage of the DC_LINK signal does not exceed TRIP_REF.Consequently, the current regulator 22 triggers an overcurrent eventbefore current in the phase winding 7 exceeds the failsafe threshold.The overcurrent module 24 and the processor 17 are thus expected toregulate current in the phase winding 7. The failsafe module 23 isexpected to clear TRIP# only in the unlikely event of a fault within theprocessor 17 (e.g. a software fault) or if the current in the phasewinding 7 rises at such a rate that the failsafe threshold, I_MAX, isreached before the processor 17 is able to respond to the overcurrentevent.

In response to an overcurrent event, the controller 16 performs adifferent series of actions depending on the speed of the rotor 5. Atspeeds below a predetermined threshold, SPEED_SINGLE, the controller 16operates in a ‘multi-switch mode’. At speeds at or above thepredetermined threshold, the controller 16 operates in a ‘single-switchmode’.

Multi-Switch Mode

In response to an overcurrent event in multi-switch mode, the controller16 freewheels the phase winding 7 by clearing FREEHWEEL#. Freewheelingcontinues for a freewheel period, T_FW, during which time current in thephase winding 7 is expected to decay to a level below the overcurrentthreshold. If current in the phase winding 7 continues to exceed theovercurrent threshold, the controller 16 again freewheels the phasewinding 7 for the freewheel period, T_FW. If, on the other hand, currentin the phase winding 7 has dropped below the overcurrent threshold, thecontroller 16 resumes excitation of the phase winding 7 by settingFREEWHEEL#.

For a particular freewheel period, T_FW, the corresponding electricalangle, A_FW, may be defined as:

A_FW(elec. deg)=T_FW(sec)*{ω(rpm)/60}*360(mech. deg)*n/2

where A_FW is the freewheel angle in electrical degrees, T_FW is thefreewheel period in seconds, ω is the rotor speed in rpm, and n is thenumber of rotor poles. Consequently, for a fixed freewheel period, thecorresponding freewheel angle increases with rotor speed. However, asthe freewheel angle increases, the remaining period over which currentand thus power is driven into the phase winding 7 decreases. Thecontroller 16 therefore employs a freewheel period, T_FW, that decreaseswith increasing rotor speed such that the corresponding freewheel angle,A_FW, does not become excessively large as the rotor 5 accelerates.

The controller 16 comprises a freewheel lookup table that stores afreewheel period for each of a plurality of rotor speeds. In response toan edge of the Z_CROSS signal, the controller 16 selects from thefreewheel lookup table a freewheel period, T_FW, corresponding to thespeed of the rotor 5. The controller 16 then uses the selected freewheelperiod to freewheel the phase winding 7 in response to an overcurrentevent. The same freewheel period, T_FW, is used by the controller 16until such time as a further edge of the Z_CROSS signal is detected. Inresponse to the further edge of the Z_CROSS signal, the controller 16selects from the freewheel lookup table a new freewheel periodcorresponding to the speed of the rotor 5. Consequently, as with theadvance period, the freewheel period is updated only when the voltage ofthe AC supply 4 crosses zero, and remains constant over each half-cycleof the AC supply 4.

The controller 16 operates in multi-switch mode only while the rotor 5accelerates from stationary to SPEED_SINGLE. As such, the length of timespent by the controller 16 in multi-switch mode is relatively short. Arelatively coarse speed resolution may therefore be used for thefreewheel lookup table without adversely affecting the power orefficiency of the motor system 1. Indeed, one might conceivably use afixed freewheel period, so long as the corresponding freewheel angledoes not become excessively large as the rotor 5 approachesSPEED_SINGLE.

At relatively low rotor speeds, the back EMF induced in the phasewinding 7 by the rotor 5 is relatively small. Consequently, current inthe phase winding 7 rises to the overcurrent threshold relativelyquickly. Owing to the relatively short period of time taken for thecurrent to reach the overcurrent threshold, the controller 16 willtypically switch the phase winding 7 between excitation and freewheelmultiple times during each electrical half-cycle of the motor 2. It isfor this reason that the controller 16 is said to operate in amulti-switch mode at speeds below SPEED_SINGLE. As the rotor speedincreases, the Hall period naturally decreases. Additionally, the backEMF increases and thus the time taken for current in the phase winding 7to reach the overcurrent threshold increases. Consequently, thecontroller 16 switches the phase winding 7 between excitation andfreewheel less frequently as the rotor 5 accelerates. Eventually, thespeed of the rotor 5 rises to a level at which the controller 16switches the phase winding 7 only once between excitation and freewheelduring each electrical half-cycle of the motor 2.

Single-Switch Mode

In response to an overcurrent event in single-switch mode, thecontroller 16 does not immediately freewheel the phase winding 7.Instead, the controller 16 continues to excite the phase winding 7 foran overrun period, T_OVR. After the overrun period has elapsed, thecontroller 16 freewheels the phase winding 7 by clearing FREEHWEEL#.Freewheeling then continues indefinitely until such time as thecontroller 16 commutates the phase winding 7. The controller 16 thusswitches the phase winding 7 from excitation to freewheel only onceduring each electrical half-cycle of the motor 2.

Referring now to FIG. 6, the overrun period, T_OVR, is defined by theequation:

T_OVR=T_OVR_OFFSET+T_OVR_AMP*abs{sin(θ)}

where T_OVR_OFFSET is an offset value, T_OVR_AMP*abs{sin(θ)} is arectified sine wave having an amplitude defined by T_OVR_AMP, and θ isthe angle in the voltage cycle of the AC supply 4.

The angle θ may be expressed as a time interval from a zero-crossing inthe voltage of the AC supply 4:

θ(deg)=t(sec)*f(Hz)* 360(deg)

where t is the time elapsed in seconds since a zero-crossing in the ACsupply 4, and f is the frequency in Hertz of the AC supply 4. Theoverrun period may then be defined as:

T_OVR=T_OVR_OFFSET+T_OVR_AMP*abs{sin(t*f*360 deg)}

More simply, the overrun period, T_OVR, may be regarded as the sum oftwo components:

T_OVR=T_OVR_OFFSET+T_OVR_SINE

where T_OVR_OFFSET is an overrun offset value that is independent oftime, and T_OVR_SINE is an overrun sine value that is dependent on time.

T_OVR_SINE is stored by the controller 16 as an overrun sine lookuptable. The overrun sine lookup table comprises an overrun sine value,T_OVR_SINE, for each of a plurality of times. In response to an edge ofthe HALL signal, the controller 16 determines the period of time, t,that has elapsed since the last edge of the Z_CROSS signal. Thecontroller 16 then selects from the overrun sine lookup table an overrunsine value, T_OVR_SINE, corresponding to the elapsed period of time. Thecontroller 16 then sums the overrun offset value, T_OVR_OFFSET, and theoverrun sine value, T_OVR_SINE, to obtain the overrun period, T_OVR.

As is described below in more detail, by selecting appropriate valuesfor the advance period, T_ADV, the overrun offset, T_OVR_OFFSET, and theoverrun amplitude, T_OVR_AMP, the efficiency of the motor system 1 mayoptimized for a specific average input power or average output power.Moreover, appropriate values may be selected such that the waveform ofcurrent drawn from the AC supply 4 complies with harmonic standards setby governing bodies.

Timeout

Irrespective of rotor speed, an overcurrent event is expected to occurat least once during each electrical half-cycle of the motor 2. Shouldan overcurrent event fail to occur, the controller 16 would continue toexcite the phase winding 7 and thus current in the phase winding 7 wouldcontinue to rise. At relatively high rotor speeds, the magnitude of backEMF in the phase winding 7 is relatively large. As a result, current inthe phase winding 7 is unlikely to reach an excessive level even in theabsence of an overcurrent event. However, at relatively low rotorspeeds, the back EMF induced in the phase winding 7 is relatively small.As a result, current in the phase winding 7 may rise to an excessivelevel in the absence of an overcurrent event. Indeed, the current mayrise to the failsafe threshold, I_MAX, which would then cause thecontroller 16 to enter Fault Mode. Consequently, when operating inmulti-switch mode, the controller 16 automatically executes theovercurrent routine after the phase winding 7 has been excitedconstantly in the same direction for a timeout period, T_TO. The timeoutperiod thus acts as a failsafe mechanism by ensuring that the maximumperiod of time over which the phase winding 7 can be excited is limited.

As the speed of the rotor 5 increases, the magnitude of back EMF inducedin the phase winding 7 also increases. Consequently, the rate at whichcurrent rises in the phase winding 7 decreases with increasing rotorspeed. Put another way, the electrical angle over which current in thephase winding 7 rises to the overcurrent threshold increases with rotorspeed. The controller 16 therefore employs a timeout angle, A_TO, thatincreases with rotor speed. For a particular timeout period, T_TO, thecorresponding timeout angle, A_TO, may be defined as:

A_TO(elec. deg)=T_TO(sec)*{ω(rpm)/60}*360(mech. deg)*n/2

where A_TO is the timeout angle in electrical degrees, T_TO is thetimeout period in seconds, ω is the rotor speed in rpm, and n is thenumber of rotor poles. Consequently, for a fixed timeout period, thecorresponding timeout angle increases linearly with rotor speed. Thecontroller 16 may therefore use a fixed timeout period, T_TO. However,better control may be achieved if the controller 16 uses a differenttimeout period for different rotor speeds. The controller 16 thereforecomprises a timeout lookup table that stores a timeout period, T_TO, foreach of a plurality of rotor speeds.

In response to an edge of the Z_CROSS signal, the controller 16 selectsfrom the timeout lookup table a timeout period, T_TO, corresponding tothe speed of the rotor 5. The same timeout period is then used by thecontroller 16 until such time as a further edge of the Z_CROSS signal isdetected. In response to the further edge of the Z_CROSS signal, thecontroller 16 selects from the timeout lookup table a new timeout periodcorresponding to the speed of the rotor 5. Accordingly, as with theadvance period and the freewheel period, the timeout period is updatedonly when the voltage of the AC supply 4 crosses zero, and remainsconstant over each half-cycle of the AC supply 4.

Constant Power

The controller 16 operates primarily in advanced commutation,single-switch mode. Within this mode, the speed of the rotor 5 varies asthe rotor 5 experiences different loads. As the rotor speed varies, sotoo does the magnitude of back EMF induced in the phase winding 7. Ifthe controller 16 were to employ a fixed advance period and overrunperiod, the average input power and the average output power of thesystem 1 would vary with rotor speed. However, there may be applicationsfor which it is desirable to have a motor system 1 that maintainsconstant average input or output power over a particular speed range.

The average input and output power of the motor system 1 are alsodependent upon the RMS voltage of the AC supply 4. However, the RMSvoltage may not be regular. Again, there may be applications for whichit is desirable to have a motor system 1 that maintains constant averageinput or output power irrespective of changes in the voltage of the ACsupply 4. Additionally, the mains power supply in two differentcountries may differ in RMS voltage but not in frequency. It would betherefore be advantageous if the same performance were achieved by themotor system 1 in both countries.

Consequently, in order to maintain constant average power (input oroutput) over a particular speed range and/or voltage range, thecontroller 16 adjusts the advance period and the overrun period inresponse to changes in rotor speed and/or RMS voltage of the AC supply4.

The advance lookup table therefore stores an advance period, T_ADV, foreach of a plurality of rotor speeds and a plurality of voltages. Thecontroller 16 also comprises an overrun offset lookup table that storesan overrun offset value, T_OVR_OFFSET, for each of a plurality of rotorspeeds and a plurality of voltages. Each lookup table is thustwo-dimensional and is indexed by rotor speed and voltage. As describedbelow in more detail, the controller 16 samples the DC_SMOOTH signal toobtain a measure of the RMS voltage of the AC supply 4, which is thenused by the controller 16 to index each of the lookup tables.

As with the advance period, the controller 16 updates the overrun offsetvalue in response to edges of the Z_CROSS signal. In particular, thecontroller 16 selects from the overrun offset lookup table an overrunoffset value, T_OVR_OFFSET, corresponding to the rotor speed and RMSvoltage of the AC supply 4. The overrun offset is thus updated only whenthe voltage of the AC supply 4 crosses zero, and remains constant overeach half-cycle of the AC supply 4.

In response to each edge of the HALL signal, the controller 16 selectsfrom the overrun sine lookup table an overrun sine value, T_OVR_SINE,corresponding to the period of time, t, that has elapsed since theprevious zero-crossing in the AC supply 4. The controller 16 then sumsthe overrun offset value, T_OVR_OFFSET, and the overrun sine value,T_OVR_SINE, to obtain the overrun period, T_OVR.

The advance period, T_ADV, and the overrun period, T_OVR, may thereforedefined as:

T_ADV=T_ADV_TABLE [speed, voltage]

T_OVR=T_OFFSET_TABLE [speed, voltage]+T_OVR_SINE_TABLE [t]

The advance period and the overrun period are thus adjusted in responseto changes in the rotor speed and the RMS voltage of the AC supply 4 soas to ensure that constant average power (input or output) is achieved.One might also adjust the overrun amplitude, T_OVR_AMP, in response tochanges in rotor speed and/or RMS voltage. For example, the controller16 might store an overrun amplitude lookup table that stores an overrunamplitude value, T_OVR_AMP, for each of a plurality of rotor speedsand/or voltages. The controller 16 would then update the overrunamplitude value in response to each edge of the Z_CROSS signal. Theoverrun sine value, T_OVR_SINE, would then be obtained by multiplyingthe overrun amplitude value, T_OVR_AMP, with the value obtained from thesine lookup table. However, the multiplication of two numbers increasesthe number of instructions executed by the controller 16. Additionally,a controller 16 having a higher bit-resolution would be needed in orderto handle the multiplication. Accordingly, in order that a relativelysimple and cheap microcontroller might be used for the controller 16,the overrun amplitude is not adjusted. Nevertheless, should it provenecessary or desirable to do so, the overrun amplitude might also beadjusted.

Constant average power is maintained over a speed range bounded bySPEED_CP_MIN and SPEED_CP_MAX, and over a voltage range bounded byV_CP_MIN and V_CP_MAX. Beyond these ranges, the controller 16 does notattempt to maintain constant average power. The reasons for this mayvary depending on the particulars of the motor system 1. For example, atvoltages below V_CP_MIN, it may not be possible to drive sufficientcurrent into the phase winding 7 over each electrical half-cycle of themotor 2 in order to maintain constant average power. Alternatively, theefficiency of the motor system 1 may drop-off significantly at voltagesbelow V_CP _MIN or maintaining constant average power below this voltagemay result in excessive current harmonics.

While constant average power is maintained over a specific speed rangeand voltage range, the motor system 1 is nevertheless able to operateeffectively at speeds and voltages beyond these ranges. The motor system1 therefore has an operating speed range defined by SPEED_MIN andSPEED_MAX, and an operating voltage range defined by V_MIN and V_MAX.The advance lookup table and the overrun offset lookup table storevalues covering the full operating speed and voltage ranges of the motorsystem 1. However, constant average power is achieved only at speedsbetween SPEED_CP_MIN and SPEED_CP_MAX, and at voltages between V_CP_MINand V_CP _MAX.

Voltage and Temperature Measurement

The peripherals 19 of the controller 16 include an analog-to-digitalconverter (ADC) having a plurality of channels. A first channel of theADC is coupled to the input pin of the DC_SMOOTH signal, and a secondchannel of the ADC is coupled to the input pin of the TEMP signal.

In order to measure the RMS voltage of the AC supply 4, the processor 17selects the first channel of the ADC and samples the DC_SMOOTH signalusing the ADC. The time constant of the R6,C2 circuit is sufficientlylong that the DC_SMOOTH signal appears relatively constant over eachhalf-cycle of the AC supply 4. The DC_SMOOTH signal thus provides ameasure of the peak voltage of the AC supply 4. Since the peak voltageis directly proportional to the RMS voltage, DC_SMOOTH also provides ameasure of the RMS voltage. Although the DC_SMOOTH signal is relativelyconstant over each half-cycle of the AC supply 4, the signalnevertheless has a small degree of high-frequency ripple that arisesfrom switching of the inverter 10. Accordingly, in order to compensatefor this ripple, the processor 17 samples the DC_SMOOTH signal numeroustimes over each cycle of the AC supply 4. The processor 17 then takesthe average of the samples in order to obtain a measure of the RMSvoltage of the AC supply 4.

In order to measure the temperature, the processor 17 selects the secondchannel of the ADC and samples the TEMP signal using the ADC. Again, theprocessor 17 samples the TEMP signal numerous times and determines theaverage in order to obtain a measure of the temperature. By taking theaverage of numerous samples, the controller 16 does not inadvertentlyreact to spurious noise in the TEMP signal or transient temperaturechanges within the motor system 1.

When operating at relatively high rotor speeds, the time required by theADC to select a channel and sample the relevant input signal may delaythe execution of other routines. As explained below in more detail, anydelay is likely to adversely affect the performance of the motor system1. Accordingly, when sampling DC_SMOOTH or TEMP, the sampling process isbroken-up into three distinct steps, each of which is executed in turnin response to an edge of the HALL signal.

In response to a first edge of the HALL signal, the processor 17 selectsthe appropriate channel of the ADC. In response to a second edge, theprocessor 17 starts the ADC. In response to a third edge, the processor17 reads the output register of the ADC. The output read by theprocessor 17 thus represents a single sample of the selected inputsignal, i.e. DC_SMOOTH or TEMP. The sample read by the processor 17 isthen stored to the memory device 18. This three-step process is thenrepeated to obtain a further sample of the input signal, which is thenadded to the value already stored in the memory device 18. The valuestored in the memory device 18 thus corresponds to the sum of theindividual samples read by the processor 17. The three-step process isrepeated a predetermined number of times. The processor 17 then dividesthe value stored in the memory device 18 by the predetermined number toobtain an average measure of the input signal.

By breaking up the sampling process into three distinct steps, the timenecessary to sample the input signal is spread across three electricalhalf-cycles of the motor 2. Consequently, the time spent by thecontroller 16 in sampling the input signal during each electricalhalf-cycle of the motor 2 is significantly reduced and thus thepossibility of event clashing is reduced.

The controller 16 samples the DC_SMOOTH and TEMP signals concurrently.As illustrated in FIG. 7, the controller 16 performs the three-stepprocess on the DC_SMOOTH signal to obtain a single sample of the RMSvoltage of the AC supply 4. The controller 16 then performs thethree-step process on the TEMP signal to obtain a single sample of thetemperature. This process of alternately sampling DC_SMOOTH and TEMP isthen repeated a predetermined number of times. The memory device 18 thusstores a first value corresponding to the sum of the samples of the RMSvoltage of the AC supply 4, and a second value corresponding to the sumof the samples of the temperature.

Rather than sampling the DC_SMOOTH and TEMP signals concurrently, thecontroller 16 may instead sample the two input signals sequentially. Inparticular, the controller 16 may sample the DC_SMOOTH signal apredetermined number of times before then sampling the TEMP signal apredetermined number of times. By sampling the two input signalssequentially rather than concurrently, the channel of the ADC is changedonly once for each set of samples. As a result, the step of selecting achannel may be dropped from all but the very first sample. A two-stepprocess (i.e. start ADC and read ADC) may then be used to sample all butthe very first of the samples. Accordingly, a greater number of samplesmay be collected over a particular time period. However, a drawback withsampling the two input signals sequentially is that, for each signal,there is a period during which the signal is not being measured.

The controller 16 has only one ADC, which is required to sample twoinput signals, namely DC_SMOOTH and TEMP. It is for this reason that thesampling process includes the step of selecting an ADC channel. If thetemperature sensor 15 were omitted from the control system 3, the RMSvoltage of the AC supply 4 could be sampled without the need for channelselection.

Alternatively, if the peripherals 19 of the controller 16 included anadditional ADC, each input signal could be sampled by a dedicated ADCand thus the step of selecting a channel could again by omitted.Nevertheless, in both instances, the sampling process continues to bebroken into two steps, such that the time necessary to sample an inputsignal is spread across two electrical half-cycles of the motor 2.

In the particular embodiment illustrated in FIG. 7, each step of thesampling process is executed in response to a successive edge of theHALL signal. This then has the advantage that each sample is obtainedrelatively quickly, i.e. after three edges of the HALL signal.Nevertheless, it is not essential that each step is executed in responseto successive edges of the HALL signal. For example, each step of thesampling process may be executed in response to every second or thirdedge of the HALL signal. While this then requires a longer period oftime to obtain each sample, the controller 16 may use the time when itis not addressing the ADC to execute other routines.

Rather than using the average of the various samples as a measure of theinput signal, the controller 16 might alternatively use the sum of thesamples. Alternatively, the controller 16 might use the peak value ofthe samples as a measure of the input signal. For example, after readinga sample from the output register of the ADC, the processor 17 mightcompare the sample against a value stored in the memory device 18. Ifthe sample is greater than the value stored in the memory device 18, theprocessor 17 overwrites the value with that of the sample. The steps ofcomparing and overwriting are then repeated for each of thepredetermined number of samples. After all samples have been collected,the value stored in the memory device 18 represents the peak value ofthe samples. When measuring the peak value, it is not essential that theinput signal representing the voltage of the AC supply 4 is smooth, solong as the samples span at least one half-cycle of the AC supply 4.Consequently, the smoothing capacitor C2 may be omitted or thecapacitance might be significantly reduced so as to reduce the sizeand/or cost of the control system 3.

In addition to measuring the voltage of the AC supply 4 and temperaturewhen operating at speed, the controller 16 also measures the voltage andtemperature during initial power up. This initial check is made in orderto ensure that the RMS voltage of the AC supply 4 and the temperaturewithin the motor system 1 lie within safe operating limits. During thisinitial stage, the time spent by the controller 16 in sampling the inputsignal is not critical. Consequently, during initial power up, thecontroller 16 samples the voltage and the temperature without breakingthe process into three steps.

Lookup Tables

The memory device 18 of the controller 16 stores a number of lookuptables, each having a particular speed and/or voltage resolution. Theresolution of each lookup table need not be the same as that of otherlookup tables and may vary across the lookup table. For example, theadvance lookup table may store an advance period every 10 krpm atrelatively low speeds, which gradually increases to 1 krpm at relativelyhigh speeds.

Advanced commutation is employed in both multi-switch mode andsingle-switch mode. In single-switch mode, the advance period isadjusted in response to changes in both rotor speed and RMS voltage ofthe AC supply 4 in order to maintain constant average power. Inmulti-switch mode, it is not necessary to adjust the advance period inresponse to changes in voltage. Accordingly, in order to minimize theamount of memory required to store the advance lookup table, the memorydevice 18 stores two advance lookup tables: a one-dimensional lookuptable that is indexed by rotor speed when operating at speeds belowSPEED_SINGLE, and a two-dimensional lookup table that is indexed byrotor speed and voltage at speeds at or above SPEED_SINGLE.

Rather than storing absolute values, each lookup table may instead storedifference values. The controller 16 then stores a reference value towhich the difference values are applied. Consequently, when updating aparticular parameter, the controller 16 indexes the relevant lookuptable to select a difference value, and applies the difference value tothe reference value to obtain the parameter. So, for example, thecontroller 16 may employ advance periods of 47 μs, 50 μs and 52 μs forthe rotor speeds 85 krpm, 90 krpm and 95 krpm. The controller 16 mightthen store 50 μs as the reference value. The advance lookup table wouldthen store −2 μs, 0 μs, and 1 μs for each of the three speeds. Storing adifference value typically requires less memory than an absolute value.As a result, the lookup tables may be stored more efficiently. A higherresolution for the lookup tables might then be achieved for a givenamount of memory. Alternatively or additionally, a cheaper controllerhaving a smaller memory capacity may be used. In a more general sense,therefore, each lookup table may be said to store a control value (e.g.an absolute or difference value) that is used by the controller 16 todetermine the relevant parameter, e.g. advance period, overrun offsetetc.

In order to reduce the number of instructions executed by the controller16, the controller 16 only updates those parameters that are requiredfor the relevant mode of operation. For example, when operating insynchronous commutation mode, the controller 16 has no need to select orupdate an advance period. Similarly, when operating in single-switchmode, the controller 16 has no need to select or update a freewheelperiod. As a consequence of updating only those parameters required fora particular mode of operation, the controller 16 does not immediatelychange from multi-switch mode to single-switch mode when the rotorreaches SPEED_SINGLE. If the controller 16 were to immediately changefrom multi-switch to single-switch mode, the controller 16 would notknow what period of time had elapsed since the previous zero-crossing inthe voltage of the AC supply 4. As a result, the controller 16 would notknow what overrun period to use. Accordingly, when the rotor speedreaches SPEED_SINGLE, the controller 16 waits until the next edge of theZ_CROSS signal before changing from multi-switch mode to single-switchmode.

As the motor 2 accelerates from stationary, the length of the Hallperiod decreases. Consequently, if a parameter (e.g. the freewheelperiod) were updated on every nth edge of the HALL signal, the intervalbetween each update would gradually decrease. Each parameter would thenbe updated less frequently at low speeds and more frequently at highspeeds. By updating each parameter in response to a zero-crossing in thevoltage of the AC supply 4, each parameter is updated at regularintervals irrespective of speed.

If a parameter were updated on every nth edge of the HALL signal whileoperating in single-switch mode, the parameter would update at differentpoints within the cycle of the AC supply 4. This in turn couldpotentially increase the harmonic content of the current waveform drawnfrom the AC supply 4. Additionally, when the motor system 1 is operatingat constant average speed, the instantaneous speed of the rotor 5nevertheless varies over each half-cycle of the AC supply 4 owing to thesinusoidal increase and decrease in the DC link voltage. If a parameterwere updated on every nth edge of the HALL signal, a different value forthe parameter might be selected in spite of the fact that the averagespeed of the motor system 1 had not changed. Again, this may result inincreased harmonics within the current waveform drawn from the AC supply4. By updating each parameter in response to a zero-crossing in the ACsupply 4, the same reference point in the cycle of the AC supply 4 isused. Consequently, a more stable current waveform is achieved.Furthermore, by updating the parameters only once every half-cycle ofthe AC supply 4, the instructions executed by the controller 16 are keptrelatively simple and thus a simpler and cheaper microcontroller may beused. Of course, if one so desired, the various parameters may beupdated less frequently by updating on every nth edge of the Z_CROSSsignal.

Motor Operation

The operation of the motor system 1 as it accelerates from stationary torunning speed will now be described. As can be seen from FIG. 8, thecontroller 16 has six operating modes: Initialization, Stationary,Low-Speed Acceleration, High-Speed Acceleration, Running, and Fault.Within the various operating modes, the controller 16 controls the motor2 through the use of one or more of the following four parameters:freewheel period, advance period, overrun period, and timeout period.

Initialization Mode

On powering up, the controller 16 enables the peripherals 19 and samplesthe DC_SMOOTH signal and the TEMP signal in order to obtain a measure ofRMS voltage of the AC supply 4 and the temperature within the motorsystem 1. If the RMS voltage is less than an under-voltage threshold,V_MIN, or greater than an over-voltage threshold, V_MAX, the controller16 writes an ‘Under Voltage’ or ‘Over Voltage’ error to the memorydevice 18 and enters Fault Mode. Similarly, if the temperature is lessthan an under-temperature threshold, TEMP_MIN, or greater than anover-temperature threshold, TEMP_MAX, the controller 16 writes an ‘UnderTemperature’ or ‘Over Temperature’ error to the memory device 18 andenters Fault Mode.

If the RMS voltage and the temperature lie within the operatingthresholds, the controller 16 determines whether the speed of the rotor5 exceeds a stationary threshold, SPEED_STATIONARY. As noted above, thespeed of the rotor 5 is obtained from the interval between twosuccessive edges of the HALL signal, i.e. the Hall period. If thecontroller 16 fails to detect two edges of the HALL signal within aperiod of time corresponding to SPEED_STATIONARY, the controller 16enters Stationary Mode. Otherwise, the controller 16 enters Low SpeedAcceleration Mode.

Stationary Mode (ω<SPEED_STATIONARY)

The controller 16 drives the motor 2 in reverse for a predeterminedreverse-drive time, T_RD. For the purposes of the present description,it will be assumed that the motor 2 is driven forwards in response todriving the phase winding 7 from left-to-right when the HALL signal islogically low and from right-to-left when the HALL signal is logicallyhigh. The motor 2 is therefore driven in reverse in response to drivingthe phase winding 7 from right-to-left when the HALL signal is logicallylow and from left-to-right when the HALL signal is logically high, asdetailed in FIG. 9.

Momentarily driving the motor 2 in reverse should cause the rotor 5 toeither rotate in a forward direction or adopt a particular angularposition relative to the stator 6. Whether the rotor 5 rotates forwardor aligns with the stator 6 will depend on the starting position of therotor 5. The rotor 5 is therefore either moving in a forward directionor is in a position ready for acceleration in a forward direction.

After momentarily driving the motor 2 in reverse, the controller 16commutates the phase winding 7 so as to drive the motor 2 forward. Theforward drive should cause the rotor 5 to rotate in a forward direction.If the rotor 5 is rotating as expected, an edge of the HALL signalshould occur within a predetermined time, T_FD. If no edge is detectedwithin the predetermined time, T_FD, the controller writes a ‘Fail toStart’ error to the memory device 18 and enters Fault Mode. Otherwise,the controller 16 commutates the phase winding 7 in response to the edgeof the HALL signal so as to continue driving the motor 2 forward. Asecond edge of the HALL signal should then occur within a period of timecorresponding to SPEED_STATIONARY. If the second edge is detected withinthe predetermined time, the controller 16 enters Low-Speed AccelerationMode. Otherwise, the controller writes a ‘Fail to Start’ error to thememory device 18 and enters Fault Mode.

Low-Speed Acceleration Mode (SPEED_STATIONARY—ω<SPEED ADV)

When operating in Low-Speed Acceleration Mode, the controller 16 drivesthe motor 2 in synchronous commutation, multi-switch mode. FIG. 10illustrates the waveforms of the HALL signal, the control signals, andthe phase current over a few Hall periods.

In response to each edge of the HALL signal, the controller 16immediately commutates the phase winding 7 (i.e. by reversing DIR1 andDIR2, and by setting FREEWHEEL#). The controller 16 then determines thespeed of the rotor 5 on the basis of the Hall period, T_HALL. Thecontroller 16 then polls a zero-cross flag, which is set in response toan edge of the Z_CROSS signal. If the zero-cross flag is set and thespeed of the rotor 5 is greater than or equal to SPEED_ADV, thecontroller 16 enters High-Speed Acceleration Mode. If, on the otherhand, the zero-cross flag is set but the rotor speed is less thanSPEED_ADV, the controller 16 updates the freewheel period, T_FW, and thetimeout period, T_TO, and clears the zero-cross flag. The freewheel andtimeout periods are updated by indexing the freewheel and timeout lookuptables using the rotor speed.

After polling the zero-cross flag, and if necessary updating thefreewheel and timeout periods, the controller 16 performs one of thethree steps used to sample the DC_SMOOTH and TEMP signals. If thepredetermined number of samples has been collected, the controller 16determines the average of the samples to obtain a measure of the RMSvoltage of the AC supply 4 or the temperature within the motor system 1.If the RMS voltage is less than V_MIN or greater than V_MAX, or if thetemperature is less than TEMP_MIN or greater than TEMP_MAX, thecontroller 16 writes a corresponding error to the memory device 18 andenters Fault Mode.

Following commutation, the controller 16 continues to excite the phasewinding 7 until either an overcurrent event occurs or the timeoutperiod, T_TO, expires. In response to either of these two events, thecontroller 16 freewheels the phase winding 7 (i.e. by clearingFREEHWEEL#) for the freewheel period, T_FW. If, at the end of thefreewheel period, current in the phase winding 7 exceeds the overcurrentthreshold, the controller 16 again freewheels the phase winding 7 forthe freewheel period, T_FW. Otherwise, at the end of the freewheelperiod, the controller 16 resumes excitation of the phase winding 7(i.e. by setting FREEWHEEL#).

The controller 16 therefore commutates the phase winding 7 in synchronywith the edges of the HALL signal, and updates the freewheel period andthe timeout period in response to edges of the Z_CROSS signal. Thecontroller 16 continues to drive the motor 2 in synchronous commutation,multi-switch mode until such time as the speed of the rotor 5 reachesSPEED_ADV. On reaching SPEED_ADV, the controller 16 enters High-SpeedAcceleration Mode in response to the next edge of the Z_CROSS signal.

High-Speed Acceleration Mode (SPEED_ADV≦ω<SPEED SINGLE)

When operating in High-Speed Acceleration Mode, the controller 16 drivesthe motor 2 in advanced commutation, multi-switch mode. FIG. 11illustrates the waveforms of the HALL signal, the control signals, andthe phase current over a few Hall periods.

In response to each edge of the HALL signal, the controller 16determines the speed of the rotor 5 on the basis of the Hall period,T_HALL. The controller 16 then polls the zero-cross flag that is set inresponse to an edge of the Z_CROSS signal. If the zero-cross flag is setand the speed of the rotor 5 is greater than or equal to SPEED_SINGLE,the controller 16 enters Running Mode. If, on the other hand, thezero-cross flag is set and the rotor speed is less than SPEED_SINGLE,the controller 16 updates the advance period, T_ADV, the freewheelperiod, T_FW, and the timeout period, T_TO, and clears the zero-crossflag. The advance, freewheel and timeout periods are updated by indexingthe corresponding lookup tables using the rotor speed.

After polling the zero-cross flag, and if necessary updating theadvance, freewheel and timeout periods, the controller 16 calculates thecommutation period, T_COM, by subtracting the advance period, T_ADV,from the Hall period, T_HALL. The controller 16 then loads thecommutation period, T_COM, into a timer.

After calculating the commutation period, the controller 16 performs oneof the three steps used to sample the DC_SMOOTH and TEMP signals. If thepredetermined number of samples has been collected, the controller 16determines the average of the samples to obtain a measure of the RMSvoltage of the AC supply 4 or the temperature within the motor system 1.If the RMS voltage is less than V_MIN or greater than V_MAX, or if thetemperature is less than TEMP_MIN or greater than TEMP_MAX, thecontroller 16 writes a corresponding error to the memory device 18 andenters Fault Mode.

The controller 16 subsequently commutates the phase winding 7 (i.e. byreversing DIR1 and DIR2, and by setting FREEWHEEL#) after the timer hascounted for the commutation period, T_COM. As a result, the controller16 commutates the phase winding 7 in advance of the next edge of theHALL signal by the advance period, T_ADV. Following commutation, thecontroller 16 excites the phase winding 7 until either an overcurrentevent occurs or the timeout period, T_TO, expires. In response to eitherof these two events, the controller 16 freewheels the phase winding 7(i.e. by clearing FREEHWEEL#) for the freewheel period, T_FW. If, at theend of the freewheel period, current in the phase winding 7 exceeds theovercurrent threshold, the controller 16 again freewheels the phasewinding 7 for the freewheel period, T_FW. Otherwise, at the end of thefreewheel period, the controller 16 resumes excitation of the phasewinding 7 (i.e. by setting FREEWHEEL#).

The controller 16 therefore commutates the phase winding 7 in advance ofthe edges of the HALL signal, and updates the advance period, thefreewheel period and the timeout period in response to edges of theZ_CROSS signal. The controller 16 continues to drive the motor 2 inadvanced commutation, multi-switch mode until such time as the speed ofthe rotor 5 reaches SPEED_SINGLE. On reaching SPEED_SINGLE, thecontroller 16 enters Running Mode in response to the next edge of theZ_CROSS signal.

Running Mode (SPEED_SINGLE≦ω)

When operating in Running Mode, the controller 16 drives the motor 2 inadvanced commutation, single-switch mode. FIG. 12 illustrates thewaveforms of the HALL signal, the control signals, and the phase currentover a few Hall periods.

In response to each edge of the HALL signal, the controller 16determines the speed of the rotor 5 on the basis of the Hall period,T_HALL. The speed of the rotor 5 is expected to remain within a speedrange bounded by SPEED_MIN and SPEED_MAX. The controller 16 will,however, permit transient speeds outside of this range. Accordingly, ifthe speed of the rotor 5 drops below SPEED_MIN for a period of timelonger than T_US, the controller 16 writes an ‘Under Speed’ error to thememory device 18 and enters Fault Mode. Similarly, if the speed of therotor 5 exceeds SPEED_MAX for a period of time longer than T_OS, thecontroller 16 writes an ‘Over Speed’ error to the memory device 18 andenters Fault Mode. However, should the speed of the rotor 5 exceedSPEED_TRIP, the controller 16 immediately writes a ‘Speed Trip’ error tothe memory device 18 and enters Fault Mode. At speeds in excess ofSPEED_TRIP, the likelihood of mechanical and/or electrical failuresignificantly increases.

The controller 16 then polls the zero-cross flag that is set in responseto an edge of the Z_CROSS signal. If the zero-cross flag is set, thecontroller 16 updates the advance period, T_ADV, and the overrun offsetvalue, T_OVR_OFFSET. Each value is updated by indexing the relevantlookup table using the rotor speed and the measured RMS voltage of theAC supply 4. After updating the advance period and the overrun offsetvalue, the controller 16 clears the zero-cross flag, and starts azero-cross timer.

After polling the zero-cross flag, and if necessary updating the advanceperiod and the overrun offset value, the controller 16 calculates thecommutation period, T_COM, by subtracting the advance period, T_ADV,from the Hall period, T_HALL. The controller 16 then loads thecommutation period, T_COM, into a timer. After determining thecommutation period, the controller 16 indexes the overrun sine lookuptable using the time stored by the zero-cross timer, t, to select anoverrun sine value, T_OVR_SINE. The controller 16 then sums the overrunoffset value, T_OVR_OFFSET, and the overrun sine value, T_OVR_SINE, toobtain the overrun period, T_OVR.

After determining the commutation and overrun periods, the controller 16performs one of the three steps used to sample the DC_SMOOTH and TEMPsignals. If the predetermined number of samples has been collected, thecontroller 16 determines the average of the samples to obtain a measureof the RMS voltage of the AC supply 4 or the temperature within themotor system 1. If the RMS voltage is less than V_MIN or greater thanV_MAX, or if the temperature is less than T_MIN or greater than T MAX,the controller 16 writes a corresponding error to the memory device 18and enters Fault Mode.

The controller 16 subsequently commutates the phase winding 7 (i.e. byreversing DIR1 and DIR2, and by setting FREEWHEEL#) after the timer hascounted for the commutation period, T_COM. As a result, the controller16 commutates the phase winding 7 in advance of the next edge of theHALL signal by the advance period, T_ADV. Following commutation, thecontroller 16 excites the phase winding 7 until an overcurrent eventoccurs. In response to the overcurrent event, the controller 16continues to excite the phase winding 7 for the overrun period, T_OVR.After the overrun period has elapsed, the controller 16 freewheels thephase winding 7 (i.e. by clearing FREEHWEEL#). Freewheeling thencontinues indefinitely until such time as the controller 16 nextcommutates the phase winding 7. The controller 16 therefore commutatesthe phase winding 7 in advance of each edge of the HALL signal, updatesthe overrun period in response to each edge of the HALL signal, andupdates the advance period and the overrun offset value in response toeach edge of the Z_CROSS signal.

When operating in Running Mode, the magnitude of back EMF induced in thephase winding 7 by the rotor 5 is of sufficient magnitude that currentin the phase winding 7 is unlikely to rise to an excessive level, evenin the absence of an overcurrent event. As a result, no timeout periodis employed by the controller 16 when operating in Running Mode. Thisthen reduces the number of instructions executed by the controller 16.

The controller 16 drives the motor 2 over an operating speed rangebounded by SPEED_MIN and SPEED_MAX, the speed varying in response tochanges in load. Within this speed range, the controller 16 selectscontrol values that ensure that constant average power is achievedbetween SPEED_CP_MIN and SPEED_CP_MAX. Consequently, constant averagepower is achieved for different loading. The controller 16 also drivesthe motor 2 over a voltage range bounded by V_MIN and V_MAX. Within thisvoltage range, the controller 16 selects control values that ensure thatconstant average power is achieved between V_CP_MIN and V_CP MAX.Consequently, the same power and performance is achieved irrespective ofvariances in the voltage of the AC supply 4.

Fault Mode

The controller 16 enters Fault Mode in response to an error, with theintention of preventing or limiting damage to the motor system 1. Thecontroller 16 therefore disables the motor 2 by clearing DIR1 and DIR2on entering Fault Mode. The controller 16 may require that power to themotor system 1 be turned off before the motor system 1 can be reused.Alternatively, the controller 16 may prevent any further use of themotor system 1; this may depend on the type of fault that has occurred.

Benefits

For a conventional permanent-magnet motor that is driven by an ACsupply, the back EMF induced in the phase winding makes it difficult toaccurately control the amount of current drawn from the AC supply. As aresult, the waveform of current drawn from the AC supply will typicallyhave a high harmonic content, resulting in a poor power factor. In orderto address this problem, conventional permanent-magnet motors generallyinclude an active power factor correction (PFC) circuit. The active PFCcircuit outputs a regular DC voltage for use in exciting the phasewindings while ensuring that current drawn from the AC supply issubstantially sinusoidal. As a result, a relatively high power factorcan be achieved. However, the inclusion of an active PFC circuitincreases the cost of the motor system. Additionally, the PFC circuitrequires a high-capacitance DC link capacitor in order that the DC linkvoltage sampled by the PFC circuit is stable. Without a stable DC linkvoltage, the PFC circuit would estimate incorrect current demand levels,resulting in poor current harmonics. However, a high-capacitance DC linkcapacitor is both physically large and expensive.

With the motor system 1 of the present invention, the controller 16employs an overcurrent threshold that is directly proportional to the DClink voltage and an overrun period that varies across each half-cycle ofthe AC supply 4. The net result is that the controller 16 excites thephase winding 7, during each electrical half-cycle of the motor 2, for aconduction period that varies across each half-cycle of the AC supply 4.In particular, the length of the conduction period varies substantiallyas a half-sinusoid over each half-cycle of the AC supply 4. As a result,the waveform of current drawn from the AC supply 4 approaches that of asinusoid. A relatively high power factor and low harmonic content aretherefore achieved without the need for a PFC circuit orhigh-capacitance link capacitor.

FIG. 13 illustrates the current waveform that is achievable with themotor system 1 of the present invention. The current waveform isoverlaid with a perfect sinusoid for the purposes of comparison. Byemploying a conduction period that varies across each half-cycle of theAC supply 4, a current waveform may be realized for which the amplitudeof low-order harmonics is relatively small. The high-frequency ripplethat can be seen in the current waveform of FIG. 13 arises from inverterswitching.

A power factor of unity is achieved for a current waveform having noharmonic content. As the harmonic content increases, the power factordecreases. The motor system 1 of the present invention is therefore ableto achieve a relatively high power factor. Indeed, with the motor system1 of the present invention, a power factor of at least 0.95 isachievable. As a result, the motor system 1 is able to achieve arelatively high average input power for a given peak current. Incontrast, a motor system having relatively large low-order harmonicswill suffer from a poor power factor. As a result, a lower average inputpower is achieved for the same peak current. In order to remedy this,the level of peak current might be increased. However, as the peakcurrent increases, the efficiency of the system decreases due toincreased power losses. Additionally, excessive peak currents may damagethe switches of the inverter and/or demagnetize the rotor.

Many countries have regulations that impose strict limits on themagnitude of the current harmonics that may be drawn from the mainspower supply, e.g. IEC61000-3-2. By employing suitable values for theadvance period, the overrun offset, and the overrun amplitude, the motorsystem 1 is able to comply with the harmonic standards across the fulloperating speed and voltage ranges of the motor system 1. Indeed, thecurrent waveform of FIG. 13, while not perfectly sinusoidal, complieswith harmonic standards set out in IEC61000-3-2.

In contrast to conventional motor systems, the motor system 1 of thepresent invention achieves a current waveform having relatively smalllow-order harmonics without the need for an active PFC circuit or ahigh-capacitance link capacitor. The link capacitor Cl of the controlsystem 3 is employed only to smooth the relatively high-frequency ripplethat results from inverter switching. The link capacitor C1 is notrequired to smooth the DC link voltage at the fundamental frequency. Assuch, a link capacitor may be used that results in ripple in the DC linkvoltage of 50% or more at the fundamental frequency, i.e.Vr=(Vmax−Vmin)/Vmax≧0.5. The controller 16 nevertheless ensures that,even at this level of ripple, low-order harmonics are kept relativelysmall and thus a good power factor can be achieved at relatively highaverage input power. Indeed, the current waveform of FIG. 13 is achievedwith a voltage ripple of 100%. Since the link capacitor Cl is requiredonly to filter high-frequency switching ripple, a relatively lowcapacitance link capacitor may be used, thus significantly reducing thecost and size of the motor system 1.

Owing to the relatively high power factor that is achievable with themotor system 1, a relatively high average input power can be achieved inspite of the ripple in the DC link voltage. The achievable average inputpower will naturally depend on the RMS voltage of the AC supply 4.However, for an RMS voltage of 100 V, a constant average input power inexcess of 1000 W is achievable irrespective of ripple in the DC linkvoltage. Consequently, when used with a mains power supply, the motorsystem 1 is able to achieve a constant average input power of at least1000 W.

By selecting appropriate values for the advance period, T_ADV, theoverrun offset, T_OVR_OFFSET, and the overrun amplitude, T_OVR_AMP, adesired average input or output power may be achieved for the motorsystem 1. Moreover, appropriate values may be selected such that theefficiency of the motor system 1 at each operating point (i.e. speed andvoltage) is optimized for the desired input or output power. That is tosay that various sets of values for T_ADV, T_OVR_OFFSET, and T_OVR_AMPmay result in the same desired average input or output power. However,from these various sets of values, a single set may be selected thatprovides the best efficiency.

One or more of the advance period, the overrun offset, and the overrunamplitude may be adjusted in response to changes in rotor speed and/orRMS voltage of the AC supply 4 such that a particular profile for theaverage input or output power is achieved over a speed range and/orvoltage range. In particular, by adjusting at least the advance periodand the overrun offset in response to changes in rotor speed and/or RMSvoltage, the same average input or output power may be achieved.

For a single motor system, appropriate values may be selected such thatthe variance in average power (input or output) does not exceed ±1% overa speed range spanning at least 10 krpm and/or a voltage range spanningat least 10 V. However, if the same values are used in a plurality ofmass-produced motor systems, the variance in average power for eachmotor system increases owing to component and manufacturing tolerances.Nevertheless, appropriate values may be selected such that the variancein average power does not exceed ±5% for a mass-produced motor systemover the aforementioned speed and voltage ranges. Constant average power(i.e. within ±5%) may also be achieved at relatively high speeds. Inparticular, constant average power may be achieved over a speed rangehaving a minimum greater than 60 kpm and a maximum greater than 80 krpm.Indeed, constant average power may be achieved at speeds in excess of100 krpm. In addition to achieving constant average power over a speedand/or voltage range, appropriate values may be selected such that anefficiency of at least 80% is maintained over the speed and/or voltagerange.

The present invention therefore provides a high-power motor system 1that is able to comply with existing harmonic standards without the needfor an active PFC circuit or high-capacitance link capacitor. Moreover,the motor system 1 is able to achieve a relatively high efficiency (i.e.at least 80%) as well as constant average power (i.e. within ±5%) over arange of rotor speeds and RMS voltages.

Event Clashing

The controller 16 executes different software routines in response todifferent events. For example, the controller 16 executes a particularroutine in response to an edge of the HALL signal. The controller 16executes a different routine in response to an overcurrent event, and soon.

A relatively simple microcontroller typically includes a single-threadedprocessor. Consequently, when the processor executes a routine inresponse to a particular event, the processor is unable to respond toother events until such time as it has finished executing the routine.Accordingly, when two events clash, the execution of one of the eventroutines will be delayed.

When operating at relatively low rotor speeds, any delay to theexecution of a particular routine will be relatively small in comparisonto the overall Hall period. Consequently, the delay is unlikely toadversely affect the performance of the motor system 1. Moreover, thetime spent at speeds below SPEED_SINGLE is expected to be relativelyshort, and thus any effect that event clashing may have on theperformance of the motor system 1 is not regarded as critical at thesespeeds. However, at speeds at or above SPEED_SINGLE, any delay in theexecution of a routine may adversely affect the performance of the motorsystem 1. In particular, the delay may affect one or more of the inputpower, output power, efficiency and current harmonics.

For example, when operating in single-switch mode, the controller 16calculates the commutation period, T_COM, and the overrun period, T_OVR,in response to each edge of the HALL signal. If during this time, anovercurrent event occurs, the overcurrent routine will not be executeduntil such time as the controller 16 has finished executing the Hallroutine. As a result, more current than ideally desired would be driveninto the phase winding 7. Alternatively, if an edge of the HALL signalwere to occur while the controller 16 is executing the overcurrentroutine, the execution of the Hall routine would be delayed. Since theHall routine is used to calculate the time at which the phase winding 7is commutated, any delay in the execution of the Hall routine will havethe effect of reducing the advance period. In each of these examples,the clash is likely to occur around the zero-crossing in the voltage ofthe AC supply 4, owing to the small overcurrent threshold. Consequently,in spite of the fact that the amount of current driven into the phasewinding 7 is not well controlled, the net effect on power and efficiencyis unlikely to be significant. However, the net effect on currentharmonics is likely to be significant.

Various measures may be taken in order to minimize the risk of eventclashing. In particular, the risk of clashing may be reduced bysimplifying the instructions of each routine such that the time requiredto execute each routine is kept relatively short. It is for this reasonthat the controller 16 uses lookup tables that store control values inthe form of time periods. By using lookup tables that store timeperiods, the mathematical calculations performed by the controller 16may be kept relatively simple. In particular, the mathematicalcalculations can be limited to simple addition (e.g. when calculatingthe overrun period) and subtraction (e.g. when calculating thecommutation period). Nevertheless, in spite of these measures, eventclashing may occur with a relatively simple processor at relatively highspeeds.

Event clashing may be resolved by having a faster or multi-coreprocessor. However, both options increase the cost of the controller 16.Accordingly, two alternative schemes for driving the motor 2 insingle-switch mode will now be described. Both schemes reduce the numberof events that occur during each electrical half-cycle of the motor 2and thus reduce the possibility of event clashing. Before describing thetwo alternative schemes, consideration will first be made of the eventsthat occur during each electrical half-cycle of the motor 2 for theabove-described scheme. For the purposes of clarity, the control schemedescribed above for single-switch mode shall hereafter be referred to as‘overcurrent single-switch mode’. The two alternative control schemesfor single-switch mode will be referred to as ‘unlimited-freewheelsingle-switch mode’ and ‘limited-freewheel single-switch mode’.

Overcurrent Single-Switch Mode

A common method of event handling is through the use of interrupts. Inresponse to an interrupt, the controller 16 interrupts execution of themain code and services the interrupt by executing an interrupt serviceroutine (ISR).

When operating in overcurrent single-switch mode, the controller 16employs the following four interrupts: Hall, overcurrent, freewheel andcommutate. FIG. 14 illustrates the waveforms of the HALL signal, thecontrol signals and the phase current, as well as the interruptsemployed by the controller 16, when operating in overcurrentsingle-switch mode.

The Hall interrupt is generated in response to an edge of the HALLsignal. In servicing the Hall interrupt, the controller 16 first pollsthe zero-cross flag that is set in response to an edge of the Z_CROSSsignal. If the zero-cross flag is set, the controller 16 updates theadvance period and the overrun offset value, and clears the zero-crossflag. The Z_CROSS signal is thus used to set a flag rather than generatean interrupt. This then minimizes the total number of interrupts andthus the possibility of interrupt clashing. After polling the zero-crossflag, the controller 16 calculates the commutation period, T_COM, andthe overrun period, T_OVR. The controller 16 then loads the commutationperiod into a first timer, Timer1. Finally, the controller 16 performsone of the three steps used to sample the DC_SMOOTH and TEMP signals.

The overcurrent interrupt is generated in response to a logically lowovercurrent signal output by the current regulator 22. In servicing theovercurrent interrupt, the controller 16 loads the overrun period,T_OVR, into a second timer, Timer2.

The freewheel interrupt is generated by the second timer when theoverrun period has elapsed. In servicing the freewheel interrupt, thecontroller 16 freewheels the phase winding 7.

The commutate interrupt is generated by the first timer when thecommutation period has elapsed. In servicing the commutate interrupt,the controller 16 commutates the phase winding 7.

Since the overcurrent ISR is responsible for loading the second timerwith the overrun period, it is not possible for the overcurrent andfreewheel interrupts to clash. Furthermore, by ensuring that the advanceperiod is longer than the time required to execute the commutate ISR,clashing of the Hall and commutate interrupts can be avoided.Nevertheless, four possible interrupt clashes are still possible, namelyHall and overcurrent, Hall and freewheel, commutate and overcurrent, andcommutate and freewheel.

Unlimited-Freewheel Single-Switch Mode

When operating in unlimited-freewheel single-switch mode, theovercurrent interrupt is disabled, i.e. the controller 16 ignores theovercurrent signal output by the current regulator 22. In response to anedge of the HALL signal, the controller 16 calculates a conductionperiod, T_CD, in addition to the commutation period, T_COM. Thecontroller 16 commutates the phase winding 7 at a time, T_COM, after theedge of the HALL signal. Following commutation, the controller 16excites the phase winding 7 for the conduction period, T_CD, after whichthe controller 16 freewheels the phase winding 7.

The conduction period resembles the overrun period employed inovercurrent single-switch mode. In particular, the conduction periodcomprises an offset value and a sine value. However, unlike the overrunperiod, the waveform of the conduction period includes a phase shiftrelative to the voltage cycle of the AC supply 4.

In overcurrent single-switch mode, the controller 16 initially excitesthe phase winding 7 until current in the phase winding 7 reaches anovercurrent threshold. Thereafter, the controller 16 excites the phasewinding 7 for the overrun period, T_OVR. The total conduction periodover which the phase winding 7 is excited is therefore the sum of theinitial excitation period and the overrun period. Current in the phasewinding 7 is sensed by monitoring the voltage across each of the shuntresistors R1,R2. More specifically, the voltage across each shuntresistors R1,R2 is output to the controller 16 as a current sensesignal, I_SENSE_1 and I_SENSE_2. As illustrated in FIG. 2, each currentsense signal is filtered by an RC filter R8,C3 and R9,C4, which acts toremove high-frequency noise. The time constant of the RC filterintroduces a time delay between the measured current and the actualcurrent in the phase winding 7. The net result is that the waveform ofthe conduction period is phase-shifted relative to the cycle of the ACsupply 4. This phase shift helps reduce the magnitude of low-ordercurrent harmonics.

In unlimited freewheel single-switch mode, the overcurrent interrupt isdisabled. The RC filters R8,C3 and R9,C4 do not therefore influence thewaveform of the conduction period. Accordingly, in order to replicatethe phase shift that exists in overcurrent single-switch mode, thewaveform of the conduction period includes a phase shift relative to thecycle of the AC supply 4. The conduction period, T_CD, is thereforedefined by the equation:

T_CD=T_CD_OFFSET+T_CD_AMP*abs{sin (θ+A_CD_PHASE)}

where T_CD_OFFSET is an offset value and T_CD_AMP*abs{sin(θ+A_CD_PHASE)} is a rectified sine wave having an amplitude defined byT_CD_AMP. θ is the angle in the voltage cycle of the AC supply 4 andA_CD_PHASE is a phase angle.Both the angle θ and the conduction phase angle, A_CD_PHASE, may beexpressed as time intervals:

θ(deg)=t(sec)*f(Hz)*360(deg) A_CD_PHASE (deg)=T_CD_PHASE(sec)*f(Hz)*360(deg)

Consequently, the conduction period may be defined as:

T_CD=T_CD_OFFSET+T_CD_AMP*abs{(sin({t+T_CD_PHASE)*f*360 deg)}

More simply, the conduction period, T_CD, may be regarded as:

T_CD=T_CD_OFFSET+T_CD_SINE

where T_CD_OFFSET is a conduction offset value that is independent oftime, and T_CD_SINE_SINE is a conduction sine value that is dependent ontime.T_CD_SINE is stored by the controller 16 as a conduction sine lookuptable, which comprises a conduction sine value, T_CD_SINE, for each of aplurality of times.

In overcurrent single-switch mode, the controller 16 adjusts the advanceperiod and the overrun offset value in response to changes in rotorspeed and voltage of the AC supply 4 so as to maintain constant averagepower. Likewise, in unlimited freewheel single-switch mode, thecontroller 16 adjusts the advance period, T_ADV, and the conductionoffset value, T_CD_OFFSET, in response to changes in rotor speed andvoltage in order to maintain constant average power. The controller 16therefore stores an advance lookup table and a conduction offset lookuptable, each of which is indexed by rotor speed and voltage:

T_ADV=T_ADV_TABLE [speed,voltage]T_CD_OFFSET=T_CD_OFFSET_TABLE[speed,voltage]

In overcurrent single-switch mode, the controller 16 initially excitesthe phase winding 7 until current in the phase winding 7 reaches anovercurrent threshold. The overcurrent threshold is proportional to theDC link voltage, and thus the length of this initial excitation periodis sensitive to changes in the voltage of the AC supply 4. The length ofthe initial excitation period is also sensitive to changes in themagnitude of the back EMF induced the phase winding 7 by the rotor 5.Consequently, the initial excitation period is sensitive to changes inboth the rotor speed and the voltage of the AC supply 4. Owing to the RCfilter that operates on each of the current sense signals, this initialexcitation period introduces a phase delay in the waveform of theconduction period, which helps reduce the magnitude of low-order currentharmonics. In unlimited-freewheel single-switch mode, the phase delay isreplicated by phase shifting the waveform of the conduction periodrelative to the voltage waveform of the AC supply 4. Since the phasedelay is sensitive to changes in the rotor speed and the voltage of theAC supply 4, the controller 16 adjusts the phase of the conductionperiod waveform in response to changes in rotor speed and voltage. Thecontroller 16 therefore comprises a conduction phase-shift lookup tablethat stores a phase-shift value, T_CD_PHASE_SHIFT, for each of aplurality of rotor speeds and a plurality of voltages. The conductionperiod may thus be defined as:

T_CD=T_CD_OFFSET_TABLE [speed,voltage]+T_CD_SINE_TABLE[t+T_CD_PHASE_SHIFT [speed,voltage]]

The advance period, T_ADV, the conduction offset value, T_CD_OFFSET, andthe conduction phase-shift value, T_CD_PHASE_SHIFT, are each updated inresponse to edges of the Z_CROSS signal. The values are thus updatedonly when the voltage of the AC supply 4 crosses zero and remainconstant over each half-cycle of the AC supply 4.

In response to an edge of the HALL signal, the controller 16 determinesthe period of time, t, that has elapsed since the last edge of theZ_CROSS signal. The controller 16 then indexes the conduction sinelookup table using the sum of the elapsed time, t, and the conductionphase shift value, T_CD_PHASE_SHIFT, in order to select a conductionsine value, T_CD_SINE. The controller 16 then sums the conduction offsetvalue, T_CD_OFFSET, and the conduction sine value, T_CD_SINE, to obtainthe conduction period, T_CD.

The controller 16 employs three interrupts when operating inunlimited-freewheel single-switch mode: Hall, freewheel and commutate.FIG. 15 illustrates the waveforms of the HALL signal, the controlsignals and the phase current, as well as the interrupts employed by thecontroller 16, when operating in unlimited-freewheel single switch mode.

The Hall interrupt is generated in response to an edge of the HALLsignal. In servicing the Hall interrupt, the controller 16 polls thezero-cross flag that is set in response to an edge of the Z_CROSSsignal. If the zero-cross flag is set, the controller 16 updates theadvance period, the conduction offset value and the conductionphase-shift value, and clears the zero-cross flag. After polling thezero-cross flag, the controller 16 calculates the commutation period,T_COM, and the conduction period, T_CD. The controller 16 then loads thecommutation period into a first timer, Timer1. Finally, the controller16 performs one of the three steps used to sample the DC_SMOOTH and TEMPsignals.

The commutate interrupt is generated by the first timer when thecommutation period has elapsed. In servicing the commutate interrupt,the controller 16 commutates the phase winding 7 and loads theconduction period into a second timer, Timer2.

The freewheel interrupt is generated by the second timer when theconduction period has elapsed. In servicing the freewheel interrupt, thecontroller 16 freewheels the phase winding 7.

In comparison to overcurrent single-switch mode, the controller 16employs one less interrupt. Moreover, since the commutate ISR isresponsible for loading the second timer with the conduction period, itis not possible for the commutation and freewheel interrupts to clash.Consequently, the risk of interrupt clashing is significantly reduced.

By ensuring that the advance period is greater than the time required toservice the commutate interrupt (T_COM _ISR) and smaller than the Hallperiod minus the time required to service the Hall interrupt(T_HALL_ISR), clashing of the Hall and commutate interrupts can beavoided, i.e.

T_COM_ISR<T_ADV<T_HALL−T_HALL_ISR

Nevertheless, it is still possible for the freewheel interrupt to clashwith the Hall interrupt. However, as will now be described, thecontroller 16 may be configured such that freewheeling is started at theend of the conduction period without the need to generate an interrupt.

Microcontrollers having a timer that is able to operate in outputcompare mode are known. In output compare mode, a comparator comparesthe counter register of the timer against an output compare register.When the values of the two registers correspond, the comparatorgenerates an interrupt or sets/clears/toggles an output pin of themicrocontroller. The particular action undertaken by the comparator istypically set by means of a register bit.

In one embodiment, output compare mode is exploited by the controller 16to clear the FREEWHEEL# signal without generating an interrupt. Asillustrated in FIG. 16, the peripherals 19 of the controller 16 comprisetwo timers 30,31 and a comparator module 32. The memory device 18comprises a timer register 33,34 for each of the timers 30,31, and acompare register 35. The first timer 30 is used to time the commutationperiod, T_COM, and the second timer 31 is used to time the conductionperiod, T_CD. The second timer 31 is configured to operate in outputcompare mode. Consequently, when servicing the commutate interruptgenerated by the first timer 30, the controller 16 commutates the phasewinding 7, loads the conduction period into the compare register 35, andresets the second timer 31. The comparator module 32 then compares thesecond timer register 34 and the compare register 35. When the tworegisters 34, 35 correspond (which occurs when the conduction period haselapsed), the comparator module 32 resets an SR latch 36, which in turnclears an output pin 21 of the controller 16. This output pin 21 is thenused by the controller 16 for the FREEWHEEL# signal. Accordingly, whenthe conduction period elapses, FREEWHEEL# is cleared without aninterrupt being generated. Since the output pin is latched, FREEWHEEL#continues to cleared until such time as the controller 16 sets the latch36 when servicing the commutate interrupt.

The controller 16 is therefore able to operate in unlimited freewheelsingle-switch mode using only two interrupts, namely Hall and commutate.However, as noted above, clashing of the Hall and commutate interruptsmay be avoided by ensuring that the advance period is kept withinparticular limits. Consequently, interrupt clashing may be avoidedaltogether.

Depending on the type of microcontroller that is used for the controller16, it may not be possible to use output compare mode to time theconduction period and clear the output pin used for FREEWHEEL#. Forexample, the microcontroller may not have any timers capable ofoperating in output compare mode. Alternatively, the microcontroller mayhave an 8-bit timer and a 16-bit timer, and only the 16-bit timer iscapable of operating in output compare mode. However, since thecommutation period is typically longer than the conduction period, itmay be necessary to use the 16-bit timer for the commutation period. Inthose instances for which output compare mode is unavailable forclearing FREEWHEEL#, a PWM module may instead be used to clearFREEWHEEL# without recourse to an interrupt, as will now be described.

FIG. 17 illustrates an alternative embodiment in which the peripherals19 of the controller 16 comprise two timers 30,31 and a PWM module 37.The memory device 18 comprises a timer register 33, 34 for each of thetimers 30,31, a duty-cycle register 38 and a period register 39. Thefirst timer 30 is again used to time the commutation period, TCOM. Thesecond timer 31, however, is used as a clock signal for the PWM module37. The PWM module 37 comprises a pair of comparators 40,41 and an SRlatch 42. A first comparator 40 compares the second timer register 34and the duty-cycle register 38. When the values of the two registers34,38 correspond, the first comparator 40 resets the SR latch 42, whichin turn clears an output pin 21 of the controller 16. A secondcomparator 41 compares the second timer register 34 and the periodregister 39. When the values of these two registers 34,39 correspond,two things happen. First, the second comparator 41 sets the SR latch 42,which in turn sets the output pin 21. Second, the second timer 31 isreset. The output pin 21 of the controller 16 is therefore cleared whenthe second timer register 34 and the duty-cycle register 38 correspond,and is set when the second timer register 34 and the period register 39correspond.

The output pin 21 toggled by the PWM module 37 is used by the controller16 for the FREEHWEEL# signal. When servicing the commutate interrupt,the controller 16 commutates the phase winding 7, loads the conductionperiod into the duty-cycle register 38, and loads the second timerregister 34 with the same value as that of the period register 39. Inresponse, the PWM module 37 sets FREEWHEEL# and the second timerregister 34 is reset. The second timer 31 then increments the secondtimer register 34 until such time as the second timer register 34 andthe duty-cycle register 38 correspond. When the two registers 34,38correspond (which occurs when the conduction period has elapsed), thePWM module 37 clears FREEWHEEL#. The phase winding 7 is thereforefreewheeled without the need for an interrupt.

If the period register 39 is set too low, the second timer register 34and the period register 39 may correspond during the freewheel period.This would then cause the FREEWHEEL# signal to be set prematurely.Accordingly, in order that freewheeling is not terminated prior tocommutation, the period register 39 stores a maximum possible value(e.g. an 8-bit period register stores 0xFF).

In each of the above embodiments, a comparator compares the value of thesecond timer register against a value stored in a compare register. Whenthe second timer register and the compare register correspond, thecomparator clears the output pin used for FREEWHEEL#. In the firstembodiment the comparator forms part of a comparator module 32, while inthe second embodiment the comparator forms part of a PWM module 37.However, any comparator of the controller 16 may be used, so long as thecomparator is able to control (either directly or through associatedhardware) an output pin of the controller 16 in response to a comparisonof a timer register and a compare register.

Limited-Freewheel Single-Switch Mode

As with unlimited-freewheel single-switch mode, the overcurrentinterrupt is disabled when operating in limited-freewheel single-switchmode. In response to an edge of the HALL signal, the controller 16calculates a drive-off period, T_DOFF, in addition to the commutationperiod, T_COM. The controller 16 then continues to excite the phasewinding 7 for the drive-off period, T_DOFF, after which the controller16 freewheels the phase winding 7. Freewheeling then continues until thecontroller 16 commutates the phase winding 7.

The drive-off period resembles the conduction period employed inunlimited-freewheel single-switch mode. In particular, the drive-offperiod, T_DOFF, is defined by the equation:

T_DOFF=T_DOFF_OFFSET+T_DOFF_AMP*abs{sin (θ+A_DOFF_PHASE)}

where T_DOFF_OFFSET is an offset value and T_DOFF_AMP*abs{sin(θ+A_DOFF_PHASE)} is a rectified sine wave having an amplitude definedby T_DOFF_AMP. θ is the angle in the voltage cycle of the AC supply 4and A_DOFF_PHASE is a phase angle.

Both the angle θ and the drive-off phase angle, A_DOFF_PHASE, may beexpressed as time intervals:

θ(deg)=t(sec)*f(Hz)*360 (deg) A_DOFF_PHASE (deg)=T_DOFF_PHASE(sec)*f(Hz)*360 (deg)

Consequently, the drive-off period may be defined as:

T_DOFF=T_DOFF_OFFSET+T_DOFF_AMP*abs{sin ({t+T_OVR_PHASE}*f*360 deg)}

More simply, the drive-off period, T_DOFF, may be regarded as:

T_DOFF=T_DOFF_OFFSET+T_DOFF_SINE

where T_DOFF_OFFSET is a drive-off offset value that is independent oftime, and T_DOFF SINE is an drive-off sine value that is dependent ontime.

The drive-off period, T_DOFF, is stored and updated by the controller 16in the same manner as that described above for the conduction period,T_CD. In particular, the controller 16 stores a drive-off sine lookuptable that is indexed by time, and a drive-off offset lookup table and adrive-off phase-shift lookup table that are each indexed by rotor speedand voltage of the AC supply 4. The drive-off period may thus be definedas:

T_DOFF=T_DOFF_OFFSET_TABLE [speed,voltage]+T_DOFF_SINETABLE[t+T_DOFF_PHASE_SHIFT [speed,voltage]]

The controller 16 employs three interrupts when operating inlimited-freewheel single-switch mode: Hall, freewheel and commutate.FIG. 18 illustrates the waveforms of the HALL signal, the controlsignals and the phase current, as well as the interrupts employed by thecontroller 16, when operating in limited-freewheel single switch mode.

The Hall interrupt is generated in response to an edge of the HALLsignal. In servicing the Hall interrupt, the controller 16 polls thezero-cross flag that is set in response to an edge of the Z_CROSSsignal. If the zero-cross flag is set, the controller 16 updates theadvance period, the drive-off offset value and the drive-off phase-shiftvalue, and clears the zero-cross flag. After polling the zero-crossflag, the controller 16 calculates the commutation period, T_COM, andthe drive-off period, T_DOFF. The controller 16 then loads thecommutation period into a first timer, Timer1, and the drive-off periodinto a second timer, Timer2. Finally, the controller 16 performs one ofthe three steps used to sample the DC_SMOOTH and TEMP signals.

The freewheel interrupt is generated by the second timer when thedrive-off period has elapsed. In servicing the freewheel interrupt, thecontroller 16 freewheels the phase winding 7.

The commutate interrupt is generated by the first timer when thecommutation period has elapsed. In servicing the commutate interrupt,the controller 16 commutates the phase winding 7.

Consequently, as with unlimited-freewheel single-switch mode, thecontroller 16 employs only three interrupts. This is in contrast to thefour interrupts used in overcurrent single-switch mode.

Moreover, by ensuring that the advance period is greater than the timerequired to service the commutate interrupt, clashing of the Hall andcommutate interrupts can be avoided.

As described above for unlimited freewheel single-switch mode, thecontroller 16 may be configured such that freewheeling is started at theend of the drive-off period without the need to generate an interrupt.For example, the second timer may be configured to operate in outputcompare mode such that the output pin used for FREEWHEEL# is clearedwhen the drive-off period elapses. Alternatively, the controller 16 mayinclude a PWM module, which is used to toggle the output pin forFREEWHEEL#. For example, when servicing the Hall interrupt, thecontroller 16 might load the commutation period into the first timer,load the drive-off period into the duty-cycle register, and reset thesecond timer. When subsequently servicing the commutate interrupt, thecontroller 16 then commutates the phase winding 7 and loads the counterregister of the second timer with the value of the period register.

The controller 16 may therefore be configured to operate in limitedfreewheel single-switch mode using only two interrupts, namely Hall andcommutate. However, as already noted, the clashing of these twointerrupts can be avoided by ensuring that the advance period is greaterthan the time required to service the commutate interrupt. Accordingly,interrupt clashing may be avoided altogether.

In limited-freewheel single-switch mode, the drive-off period isreferenced relative to an edge of the HALL signal. As a result,freewheeling of the phase winding 7 cannot commence until after the edgeof the HALL signal. In unlimited-freewheel single-switch mode, theconduction period is referenced relative to commutation. Sincecommutation occurs in advance of the edge of the HALL signal,freewheeling of the phase winding 7 may commence before, on or after theedge of the HALL signal. It is for this reason that the two schemes arereferred to as limited-freewheel and unlimited-freewheel.

In comparison to overcurrent single-switch mode, bothunlimited-freewheel and limited-freewheel single-switch modes employ asmaller number of interrupts and thus the risk of interrupt clashing isreduced. Indeed, the controller 16 may be configured such that interruptclashing is avoided altogether. Nevertheless, in spite of the potentialfor interrupt clashing, overcurrent single-switch mode has the advantageof self-compensating for tolerances and limitations within the motorsystem 1. For example, the controller 16 employs a zero-cross timer tomonitor the time elapsed since a zero-crossing in the AC supply voltage4. However, the zero-cross timer is started only as part of the Hallroutine. There is therefore a variance in the time used to index thesine lookup table. In a further example, there may be a tolerance in thebalance of the duty cycle of the HALL signal. Any imbalance in the dutycycle will introduce an error in the Hall period. Tolerances andlimitations in the motor system 1 may therefore result in small errorsin the timing of certain events (e.g. commutation, freewheeling etc). Inovercurrent single-switch mode, the controller 16 initially excites thephase winding 7 until current in the phase winding 7 reaches anovercurrent threshold. The length of this initial excitation period isnot timed by the controller 16. As a result, the initial excitationperiod acts to compensate for certain timing errors. Consequently, amore stable current waveform may be achieved when operating inovercurrent single-switch mode. In addition to self-compensating fortolerances and limitations within the motor system 1, the initialexcitation period introduces a phase delay that acts to dampen low-ordercurrent harmonics. In limited and unlimited freewheel single-switchmodes, this phase delay is replicated through the use of a phase-shiftlookup table, which consumes valuable memory resources. A cheapermicrocontroller having less memory may therefore be used for thecontroller 16 when operating in overcurrent single-switch mode.Alternatively, the memory that would otherwise be used for a phase-shiftlookup table might be used to improve the resolution of the other lookuptables, e.g. the advance, overrun offset or overrun sine lookup tables.

Conduction Period

In each of the three schemes described above, the controller 16 excitesthe phase winding 7 for a conduction period, T_CD, over each electricalhalf-cycle of the motor 2.

In overcurrent single-switch mode, the conduction period, T_CD, may bedefined as:

T_CD=T_OC+T_OVR

where T_OC is the time taken for current in the phase winding 7 to reachthe overcurrent threshold, and T_OVR is the overrun period.Consequently, the conduction period, T_CD, may be defined as:

T_CD=T_OC+T_OVR_OFFSET+T_OVRSINE

The overcurrent threshold is directly proportional to the DC linkvoltage and thus varies as a rectified sinusoid. Current in the phasewinding 7 rises substantially at the same rate irrespective of the levelof DC link voltage; the reasons for this behaviour are beyond the scopeof this document. Consequently, the time taken for current in the phasewinding 7 to reach the overcurrent threshold, T_OC, varies substantiallyas a half-sinusoid over each half-cycle of the AC supply 4. However,owing to the time constant of the RC filter that acts on each of thecurrent sense signals, the T_OC waveform is phase-shifted relative tothe voltage waveform of the AC supply 4.

The overrun offset, T_OVR_OFFSET, is constant while the overrun sinevalue, T_OVR_SINE, varies as a half-sinusoid over each half-cycle of theAC supply 4. Additionally, the waveform of the overrun sine value is inphase with the voltage waveform of the AC supply 4.

Since the overrun offset is constant over each half-cycle of the ACsupply 4, the variation in the conduction period is defined by the sumof two half-sine components, T_OC and T_OVR_SINE.

The phase difference between the two components, which arises from theRC filter, is relatively small. Additionally, the amplitude of T_OC isgreater than that of T_OVR_SINE. Consequently, in spite of the phasedifference, the sum of the two components resembles a rectified sinusoidhaving a phase shift relative to the voltage waveform of the AC supply4.

The length of the conduction period, T_CD, therefore varies as aperiodic waveform. The waveform may be defined as the sum of twocomponents: a first component (T_OVR_OFFSET) that is constant over eachcycle of the waveform and a second component that varies(T_OC+T_OVR_SINE) over each cycle of the waveform. Each cycle of thewaveform repeats with each half-cycle of the AC supply 4. However, thewaveform of the conduction period is phase-shifted relative to thevoltage waveform of the AC supply 4.

In unlimited-freewheel single-switch mode, the conduction period, T_CD,is defined by:

T_CD=T_CD_OFFSET+T_CD_SINE

The conduction offset value, TCD_OFFSET, is constant while theconduction sine value, T_CD_SINE, varies as a half-sinusoid over eachhalf-cycle of the AC supply 4. Moreover, the waveform of the conductionsine value is phase-shifted relative to the voltage waveform of the ACsupply 4. Indeed, the phase shift is intended to replicate the phaseshift that arises from the RC filter in overcurrent single-switch mode.

Consequently, as in overcurrent single-switch mode, the length of theconduction period varies as a periodic waveform. The waveform may againbe defined as the sum of two components: a first component (T_CD_OFFSET)that is constant over each cycle of the waveform and a second component(T_CD_SINE) that varies over each cycle of the waveform. Each cycle ofthe waveform repeats with each half-cycle of the AC supply 4, and thewaveform is phase-shifted relative to the voltage waveform of the ACsupply 4.

In limited-freewheel single-switch mode, the conduction period, T_CD,may be defined as:

T_CD=T_ADV+T_DOFF

where T_ADV is the advance period and T_DOFF is the drive-off period.Consequently, the conduction period, T_CD, may be defined as:

T_CD=T_ADV+T_DOFF_OFFSET+T_DOFF_SINE

The advance period, T_ADV, and the drive-off offset, T_DOFF_OFFSET, areconstant while the drive-off sine value, T_DOFF_SINE, varies as ahalf-sinusoid over each half-cycle of the AC supply 4. Again, in orderto reflect the phase shift that arises from the RC filter in overcurrentsingle-switch mode, the waveform of the drive-off sine value isphase-shifted relative to the voltage waveform of the AC supply 4.

Consequently, as with the other two single-switch modes, the length ofthe conduction period varies as a periodic waveform. The waveform may bedefined by the sum of two components: a first component(T_ADV+A_DOFF_OFFSET) that is constant over each cycle of the waveformand a second component (T_DOFF_SINE) that varies over each cycle of thewaveform. Again, each cycle of the waveform repeats with each half-cycleof the AC supply 4, and the waveform of the conduction period isphase-shifted relative to the voltage waveform of the AC supply 4.

In each of the three schemes, the length of the conduction period isdefined by a periodic waveform that repeats with each half-cycle of theAC supply 4. More particularly, the waveform varies substantially as ahalf-sinusoid over each cycle of the waveform. Consequently, thebenefits described above in connection with overcurrent single-switchmode apply equally to unlimited and limited single-switch modes.

In each scheme the waveform of the conduction period is adjusted inresponse to changes in the speed of the rotor 5 and/or the RMS voltageof the AC supply 4 so as to achieve a particular performance. Forexample, the offset of the waveform is adjusted primarily so thatconstant average power (or a particular profile for average power) isachieved over a range of speeds and/or voltages. The phase of thewaveform is adjusted primarily so that the magnitude of low-orderharmonics in the current waveform are kept below predeterminedthresholds. The length of the conduction period may be expressed as thesum of two components: a first component that is constant over eachcycle of the waveform and a second component that varies over each cycleof the waveform. In response to changes in rotor speed and/or RMSvoltage, the first component is adjusted so as to maintain constantaverage power and the second component is adjusted so as to maintainrelatively small low-order harmonics.

Although three different schemes for single-switch mode have beendescribed, the controller 16 is not necessarily limited to just one ofthese schemes when operating in single-switch mode.

Instead, the controller 16 may use one or more of the three schemes whenoperating in single-switch mode. For example, the controller 16 mayinitially employ overcurrent single-switch mode when the rotor speedreaches SPEED_SINGLE. As already noted, overcurrent single-switch modehas the benefit of providing a degree of self-compensation. However, asthe rotor 5 accelerates, the Hall period shortens and thus the risk ofinterrupt clashing increases.

Consequently, when the rotor speed reaches a predetermined threshold,the controller 16 may switch from overcurrent single-switch mode tounlimited-freewheel single-switch mode.

For each of the three single-switch schemes, values for the advanceperiod, the offset, the amplitude and the phase shift are obtained fromsimulation. The simulation refines the various values for each operatingpoint (e.g. speed and voltage) so as to obtain the best performance(e.g. best efficiency and/or low-order harmonics) at the desired averageinput or output power.

Specific Example

A particular embodiment of the motor system 1 will now be described byway of example only. Values for various hardware components of the motorsystem 1 are detailed in FIG. 19, while FIG. 20 lists various constantsand thresholds employed by the controller 16. FIGS. 21 and 22 detail theflux-linkage characteristics of the link inductor L1 and the motor 2.

As illustrated in FIG. 23, the motor system 1 has seven modes ofoperation: Fault, Initialization, Stationary, Low-Speed Acceleration,Mid-Speed Acceleration, High-Speed Acceleration, and Running.Accordingly, in comparison to that previously described and illustratedin FIG. 8, the motor system 1 has one additional mode of operation.

Fault, Initialization, Stationary and Low-Speed Acceleration Modes areunchanged from that previously described. Mid-Speed Acceleration Modecorresponds to the previously-described High-Speed Acceleration Mode.Consequently, when operating in Mid-Speed Acceleration Mode, thecontroller 16 drives the motor 2 in advanced commutation, multi-switchmode.

When operating in High-Speed Acceleration Mode, the controller 16 drivesthe motor 2 in advanced commutation, overcurrent single-switch mode. Thelength of time spent by the motor system 1 in High-Speed AccelerationMode is relatively short. Accordingly, in order to conserve memory, thecontroller 16 does not store an overrun offset lookup table and anoverrun sine lookup table. Instead, the controller 16 stores a singleoverrun lookup table that comprises an overrun period, T_OVR, for eachof a plurality of rotor speeds. The controller 16 then updates theoverrun period, along with the advance period, in response to edges ofthe Z_CROSS signal. Consequently, the overrun period employed by thecontroller 16 is constant over each half-cycle of the AC supply 4.However, the use of a constant overrun period does not adversely affectthe performance of the motor system 1 for two reasons. First, the lengthof time spent in High-Speed Acceleration mode is relatively short.Second, the controller 16 initially excites the phase winding 7 untilcurrent in the phase winding 7 exceeds a threshold that is proportionalto the DC link voltage. Consequently, in spite of using a constantoverrun period over each half-cycle of the AC supply 4, the currentwaveform continues to approach that of a sinusoid. The advance periodand the overrun period are updated in response to changes in rotor speedonly and are not updated in response to changes in the RMS voltage ofthe AC supply 4. This then reduces the size of the lookup tables, thusfreeing more memory for the more important tables used in Running Mode.The controller 16 continues to drive the motor 2 in advancedcommutation, overcurrent single-switch mode until such time as the rotorspeed reaches SPEED_UFW. On reaching SPEED_UFW, the controller 16 entersRunning Mode in response to the next edge of the Z_CROSS signal.

Running Mode corresponds to that previously described but with oneexception. Rather than employing overcurrent single-switch mode, thecontroller 16 instead employs unlimited-freewheel single-switch mode.Consequently, in addition to updating the advance period and the offsetvalue, the controller 16 also updates the phase-shift value in responseto each edge of the Z_CROSS signal. Otherwise, the operation of thecontroller 16 is essentially unchanged from that previously described.In particular, the controller 16 drives the motor 2 over an operatingspeed range bounded by SPEED_MIN and SPEED_MAX. Within this speed range,the controller 16 selects control values that ensure that constantaverage power is achieved between SPEED_CP_MIN and SPEED_CP_MAX. Thecontroller 16 also drives the motor 2 over a voltage range bounded byV_MIN and V_MAX. Within this voltage range, the controller 16 selectscontrol values that ensure that constant average power is achievedbetween V_CP_MIN and V_CP_MAX.

The controller 16 stores three advance lookup tables. The first lookuptable is a one-dimensional table that is indexed by rotor speed whenoperating in multi-switch mode. The second lookup table is aone-dimensional table that is indexed by rotor speed when operating inovercurrent single-switch mode. The third lookup table is atwo-dimensional table that is indexed by rotor speed and voltage whenoperating in unlimited-freewheel single-switch mode.

The freewheel lookup table, the timeout lookup table, and the advancelookup table used in multi-switch mode are stored collectively by thecontroller 16 as a single multi-switch map. FIG. 24 details themulti-switch map employed by the controller 16. The map stores afreewheel period, T_FW, a timeout period, T_TO, and an advance period,T_ADV, for each of a plurality of speeds. Electrical anglescorresponding to the various periods are also listed. However, theangles do not form part of the map stored by the controller 16 and areprovided merely to illustrate the behaviour of the angles with rotorspeed. For example, it can be seen that a fixed timeout period, T_TO, of70 μs is used throughout Mid-Speed Acceleration. However, thecorresponding timeout angle, A_TO, increases from 8.4 degrees at 10 krpmto 42.0 degrees at 55 krpm.

The advance and overrun lookup tables used in overcurrent single-switchmode are similarly stored as a single map. A portion of the map isdetailed in FIG. 25. Again, the corresponding electrical angles areprovided for the purposes of illustration.

The advance, conduction offset and conduction phase-shift lookup tablesused in unlimited-freewheel single-switch mode are also stored as asingle map. That is to say that the same speed and voltage resolutionare used for each of the three lookup tables. Consequently, each elementof the map stores an advance period, a conduction offset value and aconduction phase-shift value. However, for the purposes of clarity, aportion of each lookup table is illustrated in FIGS. 26-28; the unit forall three lookup tables is μs. Rather than storing absolute values, theadvance lookup table and the conduction offset lookup table storedifference values. The controller 16 then stores a reference advanceperiod of 56.2 μs and a reference conduction offset value of 48.8 μs,which correspond to a speed of 94 krpm and an RMS voltage of 230 V.

FIG. 29 details a section of the conduction sine lookup table that isstored and used by the controller 16 in unlimited-freewheel singleswitch mode. Since the frequency of the AC supply is 50 Hz, the lookuptable spans 0 to 0.01 seconds, which corresponds to a half-cycle of theAC supply 4. The resolution of the lookup table is 51.2 μsec, and theconduction amplitude, T_CD_AMP, and the conduction phase angle, T_CDPHASE are respectively 83.2 μs and 320 μs. The conduction phase angle,T_CD_PHASE, is effectively a reference phase-shift for a speed of 94krpm and a voltage of 230 V.

The motor system 1 has an operating speed range of 81 krpm to 106 krpm,and an operating voltage range of 200 V to 260 V. Within these ranges,an average input power of 1600 W 25 W is maintained at speeds between 85krpm and 106 krpm and at voltages of between 219 V to 256 V. Moreover,an efficiency of around 85% is achieved over the constant-power speedand voltage range.

The controller 16 is a PIC 16F690 microcontroller manufactured byMicrochip Technology Inc. This is a relatively simple 8-bitmicrocontroller having a clock speed of 20 MHz, a single ADC, twocomparators, three timers, 4096 words of program memory, and 512 bytesof data memory. Yet even with this relatively simple microcontroller,the controller 16 is capable of driving the motor 2 at speeds in excessof 100 krpm, with an average input power of around 1600 W.

Reference has thus far been made to a conduction period having awaveform that varies as a half-sinusoid over each cycle of the waveform(and thus over each half-cycle of the AC supply 4). However, other typesof periodic waveform may be employed for the conduction period. Inparticular, waveforms for which the conduction period varies as atriangle or trapezoid over each cycle of the waveform have both beenfound to work well in obtaining a relatively good power factor. FIG. 30illustrates the voltage waveform of the AC supply 4 along with the threeaforementioned waveforms for the conduction period: (a) half-sinusoid;(b) triangle; and (c) trapezoid. For each of these waveforms, theconduction period increases over the first half of each cycle of thewaveform and decreases over the second half of the cycle. Of these threewaveforms, the half-sinusoid has thus far been found to give the bestresults in terms of low-order harmonics. Nevertheless, for a motorsystem having different characteristics, it is quite possible thatimproved performance may be obtained using a different waveform.

In the embodiments described above, the advance period is constant overeach half-cycle of the AC supply 4. This then simplifies theinstructions executed by the controller 16. However, improvedperformance may be achieved by employing an advance period that variesacross each half-cycle of the AC supply 4.

By commutating the phase winding 7 in advance of the edges of the HALLsignal, the DC link voltage used to excite the phase winding 7 isboosted by the back EMF. As a result, the direction of current throughthe phase winding 7 may be more quickly reversed. Additionally, thecurrent in the phase winding 7 may be caused to lead the back EMF, suchthat more current may be driven into the phase winding 7 during theperiod of positive torque. As the DC link voltage increases, the timerequired to reverse the direction of phase current decreases and therate at which phase current rises increases. Accordingly, a shorteradvance period may be employed and any shortfall in the amount of phasecurrent may be made up by increasing the conduction period. Importantly,by decreasing the advance period, the period of negative torque isdecreased and thus a more efficient motor system 1 may be realized. Thecontroller 16 may therefore employ an advance period that varies acrosseach half-cycle of the AC supply 4. To this end, the length of theadvance period may be defined by a periodic waveform, each cycle of thewaveform repeating with each half-cycle of the AC supply 4. The lengthof the advance period is longer in the region around the zero-crossingsin the voltage of the AC supply 4, and is shorter in the region aroundthe peak voltage. Suitable waveforms for the advance period includeinverted half-sinusoid, inverted triangle and inverted trapezoid. FIG.31 illustrates the voltage waveform of the AC supply 4 along with threepossible waveforms for the advance period: (a) inverted half-sinusoid;(b) inverted triangle; and (c) inverted trapezoid. The advance period isdefined, stored and updated by the controller 16 in much the same manneras that described above for the conduction period. For example, anadvance period, T_ADV, that varies as an inverted half-sinusoid may bedefined as:

T_ADV=T_ADV_OFFSET−T_ADV_AMP*abs{sin ({t*f*360 deg)}

The controller 16 then uses the time that had elapsed since azero-crossing in the voltage of the AC supply 4 to determine an advanceperiod for each electrical half-cycle of the motor 2. The controller 16may additionally update the waveform of the advance period in responseto changes in rotor speed and/or changes in RMS voltage of the AC supply4. For example, the controller 16 may adjust one or more of the offset,the amplitude and the phase of the waveform in response to changes inrotor speed and/or voltage. Again, as with the conduction period, theadvance period may be defined as the sum of two components: a firstcomponent that is constant and a second component that varies acrosseach cycle of the advance period waveform. The controller 16 thenadjusts one or both components in response to changes in rotor speedand/or RMS voltage.

The parameters (e.g. advance period, conduction offset etc.) employed bythe controller 16 are adjusted in response to changes in the RMS voltageof the AC supply 4 only when operating in Running Mode. This thenreduces the size of the lookup tables that are used by the controller 16during acceleration. As a result, more memory is made available for themore important lookup tables used during Running Mode. However, theremay be instances for which it is desirable to adjust the one or moreparameters during acceleration in response to changes in the RMS voltageof the AC supply 4. For example, without adjusting the control values,the motor system 1 may start with higher power or lower power should theRMS voltage of the AC supply 4 be higher or lower than a nominatedvoltage. By adjusting the parameters during acceleration, better controlover power may be achieved. The controller 16 may therefore store avoltage compensation table for one or more of the parameters used duringacceleration, e.g. freewheel period, timeout period, advance period, andoverrun period. The voltage compensation table stores a compensationvalue for each of a plurality of voltages. When updating a particularparameter, the controller 16 indexes the relevant lookup table using therotor speed to select a control value. Additionally, the controller 16indexes the relevant voltage compensation table using the RMS voltage ofthe AC supply 4 to select a compensation value. The controller 16 thensums the control value and the voltage compensation value to obtain thevalue of the parameter. In this particular example, the voltagecompensation table is one dimensional. However, any voltage compensationideally depends not only on the RMS voltage of the AC supply 4 but alsothe speed of the rotor 5. Accordingly, rather than storing twoone-dimensional lookup tables for each parameter, the controller 16 mayinstead store a full two-dimensional lookup table for each parameter, asis done for those parameters used during Running Mode (e.g. FIG. 26-28).However, a full two-dimensional table requires considerably more memorythan two one-dimensional tables.

In the embodiments described above, the motor 2 comprises a four-polerotor 5 and a four-pole stator 6. However, the rotor 5 and stator 6might have fewer or greater number of poles. As the number of polesincreases, the number of electrical cycles per mechanical cycleincreases. Consequently, for a given rotor speed, each Hall period isshorter. A faster controller 16 might therefore be needed in order toexecute the necessary instructions during each Hall period.Additionally, faster switches for the inverter 10 might be needed. As aresult, the number of permissible poles is likely to be limited by theoperating speed of the motor 2 and/or the components of the controlsystem 3.

The current controller 22 described above and illustrated in FIG. 5makes use of the internal peripherals of the PIC16F690 microcontroller.Alternative configurations for the current controller 22 are possible,depending on the particular microcontroller that is used for thecontroller 16. Moreover, it is not essential that the current regulator22 forms part of the controller 16. Instead, the current regulator 22may be formed separately from the controller 16. The controller 16 wouldthen include an input pin 20 coupled to the current regulator 22 forreceiving the overcurrent signal.

The position sensor 13 employed by the motor system 1 is a Hall-effectsensor. However, alternative position sensors capable of outputting asignal indicative of the position of the rotor 5 might equally beemployed, e.g. optical sensor. Similarly, rather than employing a pairof clamping diodes, other arrangements may be used for the zero-crossdetector 12, e.g. Schmitt trigger.

The controller 16 freewheels the phase winding 7 by opening thehigh-side switches Q1,Q2 of the inverter 10. This then enables currentin the phase winding 7 to re-circulate around the low-side loop of theinverter 10. Conceivably, freewheeling might instead occur by openingthe low-side switches Q3,Q4 and allowing current to re-circulate aroundthe high-side loop of the inverter 10. However, the shunt resistorsR1,R2 of the current sensor 12 would then be required to be located onthe upper arms of the inverter 10 in order that current can continue tobe sensed during freewheeling. This in turn would lead to higher powerlosses since the shunt resistors R1,R2 would be subject to highervoltages during excitation. Additionally, the voltage across the shuntresistors R1,R2 would be floating rather than being referenced toneutral and thus measuring the current in the phase winding 7 would bedifficult.

In the embodiments described above, the motor system 1 comprises acontrol system 3 that drives a permanent-magnet motor 2. However, manyaspects of the control system 3 might equally be used to drive othertypes of brushless motor.

The use of a conduction period and/or an advance period that variesperiodically with time may be used to excite the phase windings of othertypes of brushless motor, e.g. reluctance motors. For a reluctancemotor, the rotor does not induce a back EMF in the phase windings of themotor. It is therefore possible to obtain a substantially sinusoidalcurrent waveform without the need for a variable conduction period oradvance period. However, a conduction period and/or an advance periodthat varies over each half-cycle of the AC supply 4 may be used toachieve a particular envelope for the magnetic flux density in themotor.

Owing to the ripple in the DC link voltage, the windings of thereluctance motor are excited with a voltage that varies over eachhalf-cycle of the AC supply 4. If a constant conduction period were usedover each half-cycle of the AC supply 4, the envelope of the magneticflux density in the motor would reflect that of the DC link voltage. Thecontroller 16 may therefore employ a conduction period that varies overeach half-cycle of the AC supply 4 so as to shape the envelope of themagnetic flux density. In particular, the controller 16 may employ aconduction period that reduces the peak magnetic flux density. Byreducing the peak magnetic flux density, a more efficient and/or asmaller motor may be realized. In order to reduce the peak magnetic fluxdensity, the controller 16 employs a conduction period that is longer inthe region around the zero-crossings in the voltage of the AC supply 4,and is shorter in the region around the peak voltage. Suitable waveformsfor the conduction period include inverted half-sinusoid, invertedtriangle and inverted trapezoid.

In order to compensate for the variation in the length of the conductionperiod, the controller 16 may additionally employ an advance period thatvaries periodically with time. In particular, as the conduction perioddecreases, the controller 16 may employ a longer advance period so as tocompensate for the shorter conduction period. Accordingly, in contrastto the conduction period, the controller 16 employs an advance periodthat is short in the region around the zero-crossings in the voltage ofthe AC supply 4, and is longer in the region around the peak voltage.Suitable waveforms for the advance period include half-sinusoid,triangle and trapezoid.

For the permanent-magnet motor 2, the controller 16 excites the phasewinding 7 in advance of zero-crossings of back EMF in the phase winding7, e.g. as determined from the signal output by the position sensor 13.For a reluctance motor, the controller 16 excites the winding in advanceof rising inductance, which may again be determined by means of aposition sensor. In both instances, the controller 16 excites the phasewinding in advance of predetermined positions of the rotor. Moreparticularly, the controller 16 excites the phase winding in advance ofunaligned rotor positions.

While a conduction period and/or an advance period that variesperiodically with time may be used with different types of brushlessmotor, variable conduction and/or advance periods are of particularbenefit when used to drive a permanent-magnet motor. As noted above, theback EMF induced in the phase winding 7 by the permanent-magnet rotor 5makes it difficult to accurately control the amount of current drawnfrom the AC supply 4. By employing a conduction period that variesperiodically with time, a waveform approaching that of a sinusoid may beachieved for current drawn from the AC power supply without the need foran active PFC or high-capacitance link capacitor.

Updating control parameters (e.g. advance period, conduction period,freewheel period and timeout period) in response to zero-crossings inthe voltage of an AC supply may be used with other types of brushlessmotor. As noted above, by updating a control parameter in response tozero-crossings in the AC supply, the control parameter is updated atregular intervals irrespective of motor speed. Moreover, the controlparameter is updated regularly without the need for a dedicated timer.The control parameter is also updated in synchrony with the cycle of theAC supply. As a result, the waveform of current drawn from the AC supplyis generally more stable.

Interrupt clashing is a potential problem for many types of brushlessmotor. Consequently, the use of a timer and comparator (e.g. formingpart of a dedicated comparator module or as part of a

PWM module) to generate a control signal in hardware rather thansoftware may be used with other types of brushless motor so as to reducethe total number of interrupts. Additionally, when the controller isrequired to sample an analog signal, interrupt clashing may be furtherreduced by dividing the sampling process into a number of steps, eachstep being performed in response to a different edge of theposition-sensor signal. As a result, the sampling process is spread overa number of electrical half-cycles of the motor, thereby freeing up moretime for the controller to execute other routines during each electricalhalf-cycle.

1. A method of controlling a brushless motor, the method comprisingexciting a winding of the motor in advance of predetermined rotorpositions by an advance period, the advance period being defined by awaveform that varies periodically with time.
 2. The method of claim 1,wherein the waveform varies substantially as one of an invertedtriangle, an inverted trapezoid and an inverted half-sinusoid over eachcycle of the waveform.
 3. The method of claim 1, wherein the methodcomprises adjusting the waveform in response to a change in one of motorspeed and excitation voltage.
 4. The method of claim 1, wherein thelength of the advance period comprises the sum of a first component anda second component, the first component is constant over each cycle ofthe waveform, and the second component varies over each cycle of thewaveform.
 5. The method of claim 4, wherein the second component variessubstantially as a half-sinusoid over each cycle of the waveform.
 6. Themethod of claim 4, wherein the method comprises adjusting the firstcomponent in response to a change in one of motor speed and excitationvoltage.
 7. The method of claim 1, wherein the method comprisesrectifying an alternating voltage to provide a rectified voltage, andexciting the winding with the rectified voltage.
 8. The method of claim7, wherein the rectified voltage has a ripple of at least 50%.
 9. Themethod of claim 7, wherein the length of the advance period varies overeach half-cycle of the alternating voltage.
 10. The method of claim 9,wherein the length of the advance period decreases over a first half ofeach half-cycle of the alternating voltage and increases over a secondhalf of each half-cycle of the alternating voltage.
 11. The method ofclaim 7, wherein the waveform of the advance period repeats with eachhalf-cycle of the alternating voltage.
 12. The method of claim 7,wherein the length of the advance period is defined by the length oftime that has elapsed since a zero-crossing in the alternating voltage.13. The method of claim 7, wherein the length of the advance periodcomprises the sum of a first component and a second component, the firstcomponent is constant over each half-cycle of the alternating voltage,and the second component varies over each half-cycle of the alternatingvoltage.
 14. The method of claim 13, wherein the method comprisesadjusting the first component in response to a change in one of motorspeed and RMS value of the alternating voltage.
 15. The method of claim13, wherein the method comprises measuring the time that has elapsedsince a zero-crossing in the alternating voltage and using the measuredtime to determine the second component.
 16. The method of claim 15,wherein the method comprises adjusting the measured time in response toa change in one of motor speed and RMS value of the alternating voltage.17. The method of claim 13, wherein the method comprises storing a firstlookup table of first control values, indexing the first lookup tableusing one of speed and voltage to select a first control value, andusing the first control value to determine the first component.
 18. Themethod of claim 13, wherein the method comprises storing a second lookuptable of second control values, measuring the time that has elapsedsince a zero-crossing in the alternating voltage, indexing the secondlookup table using the measured time to select a second control value,and using the second control value to determine the second component.19. The method of claim 18, wherein the method comprises storing a thirdlookup table of third control values, indexing the third lookup tableusing one of speed and voltage to select a third control value, andindexing the second lookup table using the measured time adjusted by thethird control value to select the second control value.
 20. A controlsystem for a brushless motor, the control system performing the methodof claim
 1. 21. The control system of claim 20, wherein the controlsystem comprises a position sensor for sensing the position of a rotorof the motor, and a controller for generating one or more controlsignals for exciting the winding in advance of the predetermined rotorpositions by the advance period, wherein the controller determines theadvance period and generates the control signals in response to eachedge of a signal output by the position sensor.
 22. The control systemof claim 20, wherein the control system comprises: a rectifier forrectifying an alternating voltage to provide a rectified voltage; azero-cross detector for detecting zero-crossings in the alternatingvoltage; an inverter coupled to the winding; and a controller forcontrolling the inverter; wherein the controller measures the time thathas elapsed since a zero-crossing in the alternating voltage, determinesthe advance period using the measured time, and generates one or morecontrol signals for exciting the winding in advance of the predeterminedrotor positions by the advance period, and the inverter excites thewinding with the rectified voltage in response to the control signals.23. A motor system comprising a permanent-magnet motor and the controlsystem of claim 20.